MIC2588, MIC2594 Datasheet by Microchip Technology

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September 2005 1 M9999-083005
MIC2588/MIC2594 Micrel
MIC2588/MIC2594
Single-Channel, Negative High-Voltage Hot
Swap Power Controllers
General Description
The MIC2588 and the MIC2594 are single-channel, nega-
tive-voltage hot swap controllers designed to address the
need for safe insertion and removal of circuit boards into
“live” high-voltage system backplanes, while using very few
external components. The MIC2588 and the MIC2594 are
each available in an 8-pin SOIC package and work in con-
junction with an external N-Channel MOSFET for which the
gate drive is controlled to provide inrush current limiting and
output voltage slew-rate control. Overcurrent fault protection
is also provided and includes a programmable overcurrent
threshold. During an output overload condition, a constant-
current regulation loop is engaged to ensure that the system
power supply maintains regulation. If a fault condition exceeds
a built-in 400µs nuisance-trip delay, the MIC2588 and the
MIC2594 will latch the circuit breaker’s output off and will
remain in the off state until reset by cycling either the UV/OFF
pin or the power to the IC. A master Power-Good signal is
provided to indicate that the output voltage of the soft-start
circuit is within its valid output range. This signal can be used
to enable one or more DC-DC converter modules.
All support documentation can be found on Micrel’s web site
at www.micrel.com.
Typical Application
/PWRGD
DRAIN
UV
OV
VDD
SENSE
GATE
VEE
1 8
7
6
54
3
2
R1
698kΩ
1%
R3
12.4kΩ
1%
R2
11.8kΩ
1%
C1
1uF
RSENSE
0.01Ω
5%
*D1
SMAT70A
100V
M1
SUM110N10-09
RFDBK
15kΩ
C3
0.22uF
CFDBK
6.8nF
100V
R4
10Ω
-48VOUT
-48V RTN
C5
100uF C4
0.1uF
-48VIN
(Long Pin)
-48V RTN
(Long Pin)
*C6
0.33uF
MIC2588-2BM
-48V RTN
(Short Pin)
DC-DC
CONVERTER
IN-
IN+
ON/OFF
#
OUT+
OUT-
+2.5V RTN
+2.5V
OUT
VDD
*C2
22nF
Nominal Undervoltage and Overvoltage Thresholds:
VUV = 36.5V
VOV = 71.2V
* Optional components (See Applications Information for more details)
# An external pull-up resistor for the power-good signal is necessary for DC-DC supplies
(and all other load modules) not equipped with internal pull-up impedence
Features
MIC2588:
Pin-for-pin functional equivalent to the
LT1640/LT1640A/LT4250
Provides safe insertion and removal from live –48V
(nominal) backplanes
Operates from –19V to –80V
Electronic circuit breaker function
Built-in 400µs “nuisance-trip” delay (tFLT)
Regulated maximum output current into faults
Programmable inrush current limiting
Fast response to short circuit conditions (< 1µs)
Programmable undervoltage and overvoltage lockouts
(MIC2588-xBM)
Programmable UVLO hysteresis (MIC2594-xBM)
Fault reporting:
Active-HIGH (-1BM) and Active-LOW (-2BM)
Power-Good output signal
Applications
Central office switching
–48V power distribution
Distributed power systems
Server Networks
Micrel, Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel + 1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
tion 1313 1313 o o CECE CECE 3313 3313 o o EEEE EEEE
MIC2588/MIC2594 Micrel
M9999-083005 2 September 2005
Pin Configuration
1PWRGD
OV
UV
VEE
8 VDD
DRAIN
GATE
SENSE
7
6
5
2
3
4
8-Pin SOIC (M)
MIC2588-1BM
1PWRGD
ON
OFF
VEE
8 VDD
DRAIN
GATE
SENSE
7
6
5
2
3
4
8-Pin SOIC (M)
MIC2594-1BM
Ordering Information
Part Number PWRGD Polarity Lockout Functions Circuit Breaker
Function
Package
Standard Pb-Free
MIC2588-1BM MIC2588-1YM Active-High Undervoltage and
Overvoltage Latched Off 8-pin SOIC
MIC2588-2BM MIC2588-2YM Active-Low Undervoltage and
Overvoltage Latched Off 8-pin SOIC
MIC2594-1BM MIC2594-1YM Active-High Programmable UVLO
Hysteresis Latched Off 8-pin SOIC
MIC2594-2BM MIC2594-2YM Active-Low Programmable UVLO
Hysteresis Latched Off 8-pin SOIC
1/PWRGD
OV
UV
VEE
8 VDD
DRAIN
GATE
SENSE
7
6
5
2
3
4
8-Pin SOIC (M)
MIC2588-2BM
1/PWRGD
ON
OFF
VEE
8 VDD
DRAIN
GATE
SENSE
7
6
5
2
3
4
8-Pin SOIC (M)
MIC2594-2BM
September 2005
September 2005 3 M9999-083005
MIC2588/MIC2594 Micrel
Pin Description
Pin Number Pin Name Pin Function
PWRGD Power-Good Output: Open-drain. Asserted when the voltage on the DRAIN
1 /PWRGD pin (VDRAIN) is within VPGTH of VEE, indicating that the output voltage is
within proper specifications.
MIC25XX-1 MIC2588-1 and MIC2594-1: PWRGD will be high-impedance when
1 PWRGD VDRAIN is less than VPGTH, and will pull-down to VDRAIN when VDRAIN is
Active-High greater than VPGTH. Asserted State: Open-Drain.
MIC25XX-2 MIC2588-2 and MIC2594-2: /PWRGD will pull-down to VDRAIN when
1 /PWRGD VDRAIN is less than VPGTH, and will be high impedance when VDRAIN is
Active-Low greater than VPGTH. Asserted State: Active-Low.
OV MIC2588: Overvoltage Threshold Input. When the voltage at the OV pin is
2 Threshold greater than the VOVH threshold, the GATE pin is immediately pulled low by
an internal 100µA current pull-down.
ON MIC2594: Turn-On Threshold. At initial system power-up or after the device
2 Turn-On Threshold has been shut off by the OFF pin, the voltage on the ON pin must exceed
the VONH threshold in order for the MIC2594 to be enabled.
UV MIC2588: Undervoltage Threshold Input. When the voltage at the UV pin is
3 Threshold less than the VUVL threshold, the GATE pin is immediately pulled low by an
internal 100µA current pull-down. The UV pin is also used to cycle the device
off and on to reset the circuit breaker. Taken together, the OV and UV pins
form a window comparator which defines the limits of VEE within which the
load may safely be powered.
OFF MIC2594: Turn-Off Threshold. When the voltage at the OFF pin is less than
3 Turn-Off Threshold the VOFFL threshold, the GATE pin is immediately pulled low by an internal
100µA current pull-down. The OFF pin is also used to cycle the device off
and on to reset the circuit breaker. Taken together, the ON and OFF pins
provide programmable hysteresis for the turn-on command voltage.
4 VEE Negative Supply Voltage Input. Connect to the negative, or low side, terminal
of the input power supply.
5 SENSE Circuit Breaker Sense Input: The current-limit threshold is set by connecting
a resistor between this pin and VEE. When the current-limit threshold of
IR = 50mV is exceeded for an internal delay tFLT (400µs), the circuit breaker
is tripped and the GATE pin is immediately pulled low by IGATEOFF. Toggling
the UV/OFF pin will reset the circuit breaker. To disable the circuit breaker,
externally connect SENSE and VEE together.
6 GATE Gate Drive Output: Connect to the gate of an external N-Channel MOSFET.
7 DRAIN Drain Sense Input: Connect to the drain of an external N-Channel MOSFET.
8 VDD Positive Supply Input. Connect to the positive, or high side, terminal of the
input power supply.
MIC2588/MIC2594 Micrel
M9999-083005 4 September 2005
Absolute Maximum Ratings(1)
(All voltages are referred to VEE)
Supply Voltage (VDD
VEE) ...........................–0.3V to 100V
DRAIN, PWRGD pins ....................................–0.3V to 100V
GATE pin ......................................................–0.3V to 12.5V
SENSE, OV, UV, ON, OFF pins .........................–0.3V to 6V
Lead Temperature (soldering)
Standard package (-xBM)
(IR Reflow, Peak Temperature) ........ 240°C +0°C/–5°C
Pb-Free Package-(xYM)
(IR Reflow, Peak Temperature) ........ 260°C +0°C/–5°C
ESD Ratings(3)
Human Body Model ................................................... 2kV
Machine Model ........................................................100V
Operating Ratings(2)
Supply Voltage (VDD
VEE) .......................... +19V to +80V
Ambient Temperature Range (TA) ................ –40°C to 85°C
Junction Temperature (TJ) ......................................... 125°C
Package Thermal Resistance
SOIC JA) ........................................................152°C/W
DC Electrical Characteristics(4)
VDD = 48V, VEE = 0V, TA = 25°C, unless otherwise noted. Bold indicates specifications apply over the full operating temperature range
of –40°C to +85°C.
Symbol Parameter Condition Min Typ Max Units
VDD – VEE Supply Voltage 19 80 V
IDD Supply Current 3 5 mA
VTRIP Circuit Breaker Trip Voltage VTRIP = VSENSE – VEE 40 50 60 mV
IGATEON GATE Pin Pull-up Current VGATE = VEE to 8V 30 45 60 µA
19V ≤ (VDD – VEE) ≤ 80V
IGATEOFF GATE Pin Sink Current (VSENSE – VEE) = 100mV 100 230 mA
VGATE = 2V
VGATE GATE Drive Voltage, (VGATE – VEE) 15V ≤ (VDD – VEE) ≤ 80V 9 10 11 V
ISENSE SENSE Pin Current VSENSE = 50mV 0.2 µA
VUVH UV Pin High Threshold Voltage Low-to-High transition 1.213 1.243 1.272 V
VUVL UV Pin Low Threshold Voltage High-to-Low transition 1.198 1.223 1.247 V
VUVHYS UV Pin Hysteresis 20 mV
VOVH OV Pin High Threshold Voltage Low-to-High transition 1.198 1.223 1.247 V
VOVL OV Pin Low Threshold Voltage High-to-Low transition 1.165 1.203 1.232 V
VOVHYS OV Pin Hysteresis 20 mV
VONH ANSI ON Pin High Threshold Low-to-High transition 1.198 1.223 1.247 V
Voltage
VOFFH ANSI OFF Pin Low Threshold High-to-Low transition 1.198 1.223 1.247 V
Voltage
ICNTRL Input Bias Current VUV = 1.25V 0.5 µA
(OV, UV, ON, OFF Pins)
VPGTH Power-Good Threshold High-to-Low transition 1.1 1.26 1.40 V
(VDRAIN – VEE)
VOLPG PWRGD Output Voltage VOLPG – VDRAIN
(relative to voltage at the DRAIN pin) 0mA ≤ IPG(LOW) ≤ 1mA
MIC25XX-1 (VDRAIN – VEE) < VPGTH –0.25 0.8 V
MIC25XX-2 (VDRAIN – VEE) > VPGTH –0.25 0.8 V
ILKG(PG) PWRGD Output Leakage Current VPWRGD = VDD = 80V 1 µA
Notes:
1. Exceeding the “Absolute Maximum Ratings” may damage the devices.
2. The devices are not guaranteed to function outside the specified operating conditions.
3. Devices are ESD sensitive. Handling precautions recommended. Human body model: 1.5kΩ in series with 100pF. Machine model: 200pF, no series
resistance.
4. Specification for packaged product only.
tr September 2005 5
September 2005 5 M9999-083005
MIC2588/MIC2594 Micrel
AC Electrical Characteristics(5)
Symbol Parameter Condition Min Typ Max Units
tFLT
Built-in Overcurrent Nuisance Trip
400 µs
Time Delay
(6)
(Figure 1)
tOCSENSE Overcurrent Sense to GATE Low VSENSE – VEE = 100mV 3.5 µs
(Figure 2)
tOVPHL OV to GATE Low(6) (Figure 3) 1 µs
tOVPLH OV to GATE High(6) (Figure 3) 1 µs
tUVPHL UV to GATE Low(6) (Figure 4) 1 µs
tUVPLH UV to GATE High(6) (Figure 4) 1 µs
tPGL(1)
DRAIN High to PWRGD Output Low(6)
RPULLUP = 100kΩ, CLOAD on PWRGD = 50pF
1 µs
(-1 Version parts only)
tPGL(2)
DRAIN Low to /PWRGD Output Low(6)
RPULLUP = 100kΩ, CLOAD on /PWRGD = 50pF
1 µs
(-2 Version parts only)
tPGH(1)
DRAIN Low to PWRGD Output High(6)
RPULLUP = 100kΩ, CLOAD on PWRGD = 50pF
2 µs
(-1 Version parts only)
tPGH(2)
DRAIN High to /PWRGD Output High(6)
RPULLUP = 100kΩ, CLOAD on /PWRGD = 50pF
2 µs
(-2 Version parts only)
Notes:
5. Specification for packaged product only.
6. Not 100% production tested. Parameters are guaranteed by design.
Timing Diagrams
I
LIMIT
I
LOAD
0A
V
DRAIN
V
GATE
(V
EE
+10V)
t < t
FLT
t t
FLT
(at VEE)
(at VEE)(at VEE)
OVERCURRENT
EVENT
Output OFF
(at VDD)
Load current is regulated
at I
LIMIT
= 50mV/R
SENSE
Reduction in VDRAIN to support
ILIMIT = 50mV/RSENSE
Figure 1. Overcurrent Response
VSENSE - VEE
100mV
1V
tOCSENSE
VGATE
Figure 2. SENSE to GATE LOW Timing Response
E E W
September 2005 6 M9999-083005
MIC2588/MIC2594 Micrel
VOV
1.223V
1V
1.203V
1V
tOVPHL
V
GATE
tOVPLH
Figure 3. Overvoltage Response
Figure 4. Undervoltage Response
V
DRAIN
MIC2588/94-1
MIC2588/94-2
V
PGTH
V
PGTH
V
EE
V
EE
V
PWRGD
Ð
V
DRAIN
= 0VV
PWRGD
Ð
V
DRAIN
= 0V
t
PGH1
VEE
VEE
PWRGD
PWRGD not asserted PWRGD not assertedPWRGD asserted - High Impedance
tPGL1
VPGTH VPGTH
tPGL2
t
PGH2
VDRAIN
/PWRGD
Figure 5. DRAIN to Power-Good Response
MIC2588/MIC2594 Micrel
M9999-083005 7 September 2005
Typical Characteristics
[Section under construction]
0
1
2
3
4
5
6
-40 -20 0 20 40 60 80 100
SUPPLY CURRENT (mA)
TEMPERATURE (°C)
Supply Current
vs. Temperature
0
1
2
3
4
5
6
15 25 35 45 55 65 75 85
SUPPLY CURRENT (mA)
SUPPLY VOLTAGE (V)
Supply Current
vs. Supply Voltage
T
A
= 25°C
0
2
4
6
8
10
12
-40 -20 0 20 40 60 80 100
VGATE (V)
TEMPERATURE (°C)
GATE Drive (VGATE - VEE)
vs. Temperature
0
2
4
6
8
10
12
10 20 30 40 50 60 70 80
VGATE (V)
SUPPLY VOLTAGE (V)
GATE Drive (VGATE - VEE)
vs. Supply Voltage
T
A
= 25°C
0
10
20
30
40
50
60
-40 -20 0 20 40 60 80 100
IGATEON (µA)
TEMPERATURE (°C)
GATE Pull-Up Current
vs. Temperature
0
50
100
150
200
250
300
350
-40 -20 0 20 40 60 80 100
IGATEOFF (mA)
TEMPERATURE (°C)
GATE Sink Current
vs. Temperature
1.2
1.21
1.22
1.23
1.24
1.25
1.26
1.27
1.28
-40 -20 0 20 40 60 80 100
UV PIN THRESHOLD (V)
TEMPERATURE (°C)
UV Pin Threshold
vs. Temperature
V
UVH
V
UVL
1
1.05
1.1
1.15
1.2
1.25
1.3
1.35
1.4
1.45
1.5
-40 -20 0 20 40 60 80 100
OV PIN THRESHOLD (V)
TEMPERATURE (°C)
OV Pin Threshold
vs. Temperature
VOVH
VOVL
1
1.05
1.1
1.15
1.2
1.25
1.3
1.35
1.4
1.45
1.5
-40 -20 0 20 40 60 80 100
POWER GOOD THRESHOLD (V)
TEMPERATURE (°C)
Power Good Threshold
vs. Temperature
VPGTH+
VPGTH–
0
0.05
0.1
0.15
0.2
0.25
0.3
0.35
0.4
-40 -20 0 20 40 60 80 100
VOLPG (V)
TEMPERATURE (°C)
Power-Good Low Voltage
vs. Temperature
1
1.05
1.1
1.15
1.2
1.25
1.3
1.35
1.4
1.45
1.5
-40 -20 0 20 40 60 80 100
VONH (V)
TEMPERATURE (°C)
ON Pin Threshold
vs. Temperature
1
1.05
1.1
1.15
1.2
1.25
1.3
1.35
1.4
1.45
1.5
-40 -20 0 20 40 60 80 100
VOFFL (V)
TEMPERATURE (°C)
OFF Pin Threshold
vs. Temperature
L J H° We?!
MIC2588/MIC2594 Micrel
M9999-083005 8 September 2005
40
42
44
46
48
50
52
54
56
58
60
-40 -20 0 20 40 60 80 100
VTRIP (mV)
TEMPERATURE (°C)
Circuit Breaker Trip Voltage
vs. Temperature
Test Circuit
/PWRGD
DRAIN
UV
OV
VDD
SENSE
GATE
VEE
1 8
7
6
54
3
2
R1
698kΩ
1%
R3
12.4kΩ
1%
R2
11.8kΩ
1%
C1
1uF
R
(see photos)
SENSE
*D1
SMAT70A
M
IRF540
1
RFDBK
10kΩ
C3
0.1uF
CFDBK
R4
10Ω
-48VOUT
-48V RTN
C5
100uF C4
0.1uF
-48VIN
(Long Pin)
-48V RTN
(Long Pin)
C2
0.22uF
MIC2588-2BM
R5
47kΩ
ILOAD
MIC2588/MIC2594 Test Circuit
MJW%WW
MIC2588/MIC2594 Micrel
M9999-083005 9 September 2005
Functional Characteristics
TIME (100µs/div.)
ILOAD
(2A/div) VSENSE
(50mV/div) VGATE
(5V/div)
Turn-On (Circuit Breaker Trip) Into Large CLOAD
RSENSE = 0.01Ω
C5 = 1500µF
CFDBK = 6.8nF
C3 = 0.22µF
Circuit Breaker Trips
TIME (25ms/div.)
ILOAD
(1A/div) DRAIN
(50V/div)
/PWRGD
(50V/div) VGATE
(5V/div)
Hot-Plug Turn-On Response
RSENSE = 0.05Ω
CFDBK = 22nF
C3 = 0.47µF
RFDBK = 1.2kΩ
TIME (100µs/div.)
ILOAD
(2A/div) DRAIN
(50V/div)
VGATE
(5V/div)
Overcurrent Response (Short Circuit)
RSENSE = 0.01Ω
CFDBK = 22nF
C2 = C3 = 0.22µF
Short Circuit Applied
TIME (5ms/div.)
ILOAD
(500mA/div) DRAIN
(20V/div)/PWRGD
(20V/div) VGATE
(10V/div)
Turn-On Response
RSENSE = 0.05Ω
CFDBK = 22nF
"H- 1O September 2005
September 2005 10 M9999-083005
MIC2588/MIC2594 Micrel
Functional Diagram
Logic +
Circuit
Breaker
Internal
PG
VEE
SENSE
VDD
+
45A
50mV
VPGTH
GATE
+
V
TH(UV/OV)
UV
+
OV
+
VDD1
VEE
Internal VDD
and
Reference
Generator
Nuisance
Trip Filter
(400s)
Current
Limit
State
VEE
EN
VEE
denotes -2 option
VEE
100A
/PWRGD
DRAIN
PWRGD
6V
Clamp
For Power Good circuitry only
V
REF1
V
DD1
V
DD1
MIC2588 Block Diagram
MIC2588/MIC2594 Micrel
M9999-083005 11 September 2005
Functional Description
Hot Swap Insertion
When circuit boards are inserted into systems carrying live
supply voltages (“hot swapped”), high inrush currents often
result due to the charging of bulk capacitance that resides
across the circuit board’s supply pins. These current spikes
can cause the system’s supply voltages to temporarily go
out of regulation, causing data loss or system lock-up. In
more extreme cases, the transients occurring during a hot
swap event may cause permanent damage to connectors or
on-board components.
The MIC2588 and the MIC2594 are designed to address these
issues by limiting the magnitude of the transient or inrush cur-
rent during hot swap events. This is achieved by controlling
the rate at which power is applied to the circuit board (di/dt
and dv/dt management). Additionally, the MIC2588 and the
MIC2594 incorporate input voltage supervisory functions and
current limiting, thereby providing robust protection for both
the system and the circuit board.
Start-Up Cycle
When the input voltage to the controller is between the over-
voltage and undervoltage thresholds (MIC2588) or is greater
than VON (MIC2594), a start cycle is initiated to deliver power
to the load. At this time, the GATE pin of the controller ap-
plies a constant charging current (IGATEON) to the gate of the
external MOSFET (M1). CFDBK creates a Miller integrator
out of the MOSFET circuit, which limits the slew-rate of the
voltage at the drain of M1. The drain voltage rate-of-change
(dv/dt) of M1 is:
dv M1
dt
I
CI
C
DRAIN GATE(–)
FDBK
GATEON
FDBK
 
where IGATE(+) = Gate Charging Current = IGATEON;
IGATE(–) –IGATE(+), due to the extremely high transconduc-
tance values of power MOSFETs; and
I C dv M1
dt
GATE(–) FDBK DRAIN
 
 
Relating the above to the maximum transient (or inrush) current
charging the load capacitance upon hot swap or power-up
involves an extension of the same formula:
IC dv M1
dt
I C I
C
| I | C
C
INRUSH LOAD DRAIN
INRUSH LOAD GATEON
FDBK
INRUSH LOAD GATEON
FDBK
 
I
(1)
The presence of C3 and RFDBK prevent turn-on of the
external pass device by limiting the hot swap current
surges induced by AC coupled transients from the drain
to the gate of M1 (i.e., CFDBK + CGD(M1)). An appropriate
value for C3 may be determined using the formula for a
capacitive voltage divider.
The maximum voltage on C3 at turn-on must be less than
VTHRESHOLD of M1.
1. For a standard 10V enhancement N-Channel
MOSFET, VTHRESHOLD is about 4.25V.
2. Choose 2V as the maximum voltage to avoid turn-
on transients.
C3 =
C
FDBK
+ C
GD
(M1)
× V
IN
(max) – V
GS
(M1)
V
GS
(M1)
V
GS
(M1) × [C3 +
C
FDBK
+ C
GD
(M1)
] = V
IN
(max) ×
C
FDBK
+ C
GD
(M1)
V
GS
(M1) × C3 =
V
IN
(max) – V
GS
(M1)

C
FDBK
+ C
GD
(M1)
(
2)
where VIN(max) = VDD – VEE(min).
For example, we can determine appropriate capacitor values
given a hot swap controller that is required to maintain the
inrush current into a 220µF load capacitance at 2A maximum
and an input supply voltage as high as VIN(max) = 75V. One
of the suggested MOSFETs to be used with the MIC2588/
MIC2594 is an SUM110N10-09,a 100V D2PAK device which
has a typical CGD of 750pF.
Calculating a value for CFBDK using Equation 1 yields:
C220 F 45 A
2A 4.95nF
FDBK  
Good engineering practice suggests the use of the worst-
case parameter values for IGATEON from the “DC Electrical
Characteristics” section:
C220 F 60 A
2A 6.6nF
FDBK  
where the nearest standard 5% value is 6.8nF. Substituting
6.8nF into Equation 2 from above yields:
C 6.8nF 750pF 75V – 2V
2V 0.275 F
3 
 
 
 
For C3, the nearest standard 5% value is 0.22µF.
While the value for RFDBK is not critical, it should be chosen
to allow a maximum of a few milliamperes to flow in the
gate-drain circuit of M1 during turn-on. While the final value
for RFDBK is determined empirically, initial values between
RFDBK = 15kΩ to 27kΩ for systems with a maximum value
of VIN(max) = 75V are appropriate.
Resistor R4, in series with the MOSFET's gate, minimizes
the potential for parasitic high frequency oscillations from
occurring in M1. While the exact value of R4 is not critical,
commonly used values for R4 range from 10Ω to 33Ω.
Power-Good (PWRGD or /PWRGD) Output
For the MIC2588-1 and the MIC2594-1, the Power-Good
output signal (PWRGD) will be high impedance when
VDRAIN drops below VPGTH, and will pull down to VDRAIN
when VDRAIN is above VPGTH. For the MIC2588-2 and the
MIC2594-2, /PWRGD will pull down to the potential of the
VDRAIN pin when VDRAIN drops below VPGTH, and will be
high impedance when VDRAIN is above VPGTH. Hence, the
-1 parts have an active-high PWRGD signal and the -2 parts
have an active-low /PWRGD output. Either PWRGD or
/PWRGD may be used as an enable signal for one or more
September 2005 12
September 2005 12 M9999-083005
MIC2588/MIC2594 Micrel
subsequent DC/DC converter modules or for other system
uses as desired. When used as an enable signal, the time
necessary for the PWRGD (or /PWRGD) signal to pull-up
(when in high impedance state) will depend upon the load
(RC) that is present on this output.
Circuit Breaker Function
The MIC2588 and the MIC2594 employ an electronic circuit
breaker that protects the MOSFET and other system compo-
nents against faults such as short circuits. The current limit
threshold is set via an external resistor, RSENSE, connected
between the VEE and SENSE pins and is determined by:
VTRIP
RSENSE
ILIM
(3)
where VTRIP is the circuit breaker trip threshold specified in
the Electrical Characteristics Table.
An internal 400µs timer limits the length of time (tFLT) for
which the circuit can draw current in excess of its programmed
threshold before the circuit breaker is tripped. This short
delay prevents nuisance tripping of the circuit breaker due
to system transients while providing rapid protection against
large-scale transient faults. Whenever the voltage across
RSENSE exceeds 50mV, two things happen:
1. A constant-current regulation loop is engaged and is
designed to hold the voltage across RSENSE equal to
50mV. This protects both the load and the MIC2588
circuit from excessively high currents. This loop will
engage in less than 1µs from the time at which the
overvoltage condition on RSENSE occurs.
2. The internal 400µs timer is started. If the 400µs
timeout period expires, the circuit breaker trips
and the GATE pin is immediately pulled low by an
internal current pull-down. This operation turns off
the MOSFET quickly and disconnects the input
from the load.
Undervoltage/Overvoltage Detection—MIC2588
The MIC2588 has “UV” and “OV” input pins. These pins can be
used to detect input supply rail undervoltage and overvoltage
conditions. Undervoltage lockout prevents energizing the load
until the supply input is stable and within tolerance. In a similar
fashion, overvoltage turn-off prevents damage to sensitive
circuit components should the input voltage exceed normal
operational limits. Each of these pins is internally connected
to an analog comparator with 20mV of hysteresis. When the
UV pin falls below its VUVL threshold or the OV pin is above
its VOVH threshold, the GATE pin is immediately pulled low.
The GATE pin will be held low until UV exceeds its VUVH
threshold or OV drops below its VOVL threshold. The UV
and OV circuits’ threshold trip points are programmed using
the resistor divider R1, R2, and R3 as shown in the “Typical
Application.” The equations to set the trip points are shown
below. For the following example, the circuit’s nominal UV
threshold is set to VUV = 37V and the nominal OV threshold
is placed at VOV = 72V, values commonly used in Central
Office power distribution applications.
V V (typ) R1+R2 +R3
R2 +R3
V V (typ) R1+R2 +R3
R3
UV UVL
OV OVH
 
 
 
 
 
(4)
(5)
Given VUV, VOV, and any one resistor value, the remaining
two resistor values can be found. A suggested value for R3
is that which will provide a minimum of 100µA of current
through the voltage divider chain at VDD = VUV. This yields
the following as a starting point:
R3 V (typ)
100 A12.23k
OVH
 
The closest standard 1% value for R3 = 12.4kΩ. Using Equa-
tions 3 and 4 above, solving for R2 and R1 yields:
R2 R3 V
V– 1
R2 12.4k 72V
37V – 1
R2 11.729k
OV
UV
 
The closest standard 1% value for R2 = 11.8kΩ. Next, the
value for R1 is calculated:
R1 R3 V –1.223V
1.223V – R2
R1 12.4k 72V – 1.223V
1.223V – 11.8kΩ
R1 705.808k
OV
 
The closest standard 1% value for R1 = 698kΩ.
Using standard 1% resistor values, the circuit’s nominal
UV and OV thresholds are:
VUV = 36.5V
VOV = 71.2V
Good general engineering design practices must consider
the tolerances associated with these parameters, including
but not limited to, power supply tolerance, undervoltgae
and overvoltage tolerances, and the tolerances of the
external passive components.
Programmable UVLO Hysteresis—MIC2594
The MIC2594 has user-programmable hysteresis by means
of the ON and OFF pins. This allows setting the part to turn
on at a voltage V1, and not turn off until a second voltage
V2, where V2 < V1. This can significantly simplify dealing
with source impedances in the supply bus while at the same
time increasing the amount of available operating time from
a loosely regulated power supply (for example, a battery
supply). Similarly to the MIC2588, each of these pins is
internally connected to an analog comparator with 20mV of
hysteresis. The MIC2594 holds the output off until the voltage
at the ON pin exceeds its VONH threshold value given in the
“Electrical Characteristics” table. Once the output has been
enabled by the ON pin, it will remain on until the voltage at
MIC2588/MIC2594 Micrel
M9999-083005 13 September 2005
the OFF pin falls below its VOFFL threshold value, or the part
turns off due to a fault. Should either event occur, the GATE
pin is immediately pulled low and will remain low until the
ON pin once again exceeds its VONH threshold. The circuit’s
turn-on and turn-off points are set using the resistor divider
R1, R2, and R3 as shown in the “Typical Application.” The
equations to establish the trip points are shown below. In the
following example, the circuit’s nominal ON threshold is set to
VON = 40V and the circuit’s nominal OFF threshold is
VOFF = 35V.
V =V (typ) R1 R2 R3
R3
V =V (typ) R1 R2 R3
R2 R3
ON ONH
OFF OFFL
 
 
 
 
 
Given VOFF, VON, and any one resistor value, the remaining
two resistor values can be readily found. A suggested value
for R3 is that which will provide a minimum of 100µA of cur-
rent through the voltage divider chain at VDD = VOFF. This
yields the following as a starting point:
R3 = V (typ)
100 A12.23k
OFFL
 
The closest standard 1% value for R3 = 12.4kΩ.
solving for R2 and R1 yields:
R2 =R3 V
V– 1
R2 =12.4k 40V
35V – 1
R2 =1.771k
ON
OFF
 
The closest standard 1% value for R2 = 1.78kΩ.
R1=R3 V –1.223V
1.223V – R2
R1=12.4k 40V – 1.223V
1.223V – 1.78kΩ
R1= 391.380k
ON
 
 
 
The closest standard 1% value for R1 = 392kΩ.
Using standard 1% resistor values, the circuit’s nominal
ON and OFF thresholds are:
VON = 40.1V
VOFF = 35V
Good general engineering design practices must consider
the tolerances associated with these parameters, including
but not limited to, power supply tolerance, undervoltgae
and overvoltage tolerances, and the tolerances of the
external passive components.
5m... Rum an Hid September 2005 14
September 2005 14 M9999-083005
MIC2588/MIC2594 Micrel
Applications Information
Optional External Circuits for Added Protection/Perfor-
mance
In many telecom applications, it is very common for circuit
boards to encounter large-scale supply-voltage transients
in backplane environments. Because backplanes present a
complex impedance environment, these transients can be
as high as 2.5 times steady-state levels, or 120V in worst-
case situations. In addition, a sudden load dump anywhere
on the circuit card can generate a very high voltage spike
at the drain of the output MOSFET which, in turn, will ap-
pear at the DRAIN pin of the MIC2588/MIC2594. In both
cases, it is good engineering practice to include protective
measures to avoid damaging sensitive ICs or the hot swap
controller from these large-scale transients. Two typical
scenarios in which large-scale transients occur are de-
scribed below:
1. An output current load dump with no bypass (charge
bucket or bulk) capacitance to VEE. For example,
if LLOAD = 5µH, VIN = 56V and tOFF = 0.7µs, the
resulting peak short-circuit current prior to the
MOSFET turning off would reach:
56V 0.7 s
5 H 7.8A
 
 
If there is no other path for this current to take
when the MOSFET turns off, it will avalanche the
drain-source junction of the MOSFET. Since the
total energy represented is small relative to the
sturdiness of modern power MOSFETs, it’s unlikely
that this will damage the transistor. However, the
actual avalanche voltage is unknown; all that can
be guaranteed is that it will be greater than the
VBD(D-S) of the MOSFET. The drain of the transistor
is connected to the DRAIN pin of the MIC2588/
MIC2594, and the resulting transient does have
enough voltage and energy to damage this, or any,
high-voltage hot swap controller.
2. If the load’s bypass capacitance (for example,
the input filter capacitors for DC-DC converter
module(s)) are on a board from which the board
with the MIC2588/MIC2594 and the MOSFET can
be unplugged, the same type of inductive transient
damage can occur to the MIC2588/MIC2594.
For many applications, the use of additional circuit compo-
nents can be implemented for optimum system performance
and/or protection. The circuit, shown in Figure 6, includes
several components to address some the following system
(dynamic) responses and/or functions: 1) suppression of
transient voltage spikes, 2) elimination of false “tripping” of the
circuit breaker due to undervoltage and overcurrent glitches,
and 3) the implementation of an external reset circuit.
It is not mandatory that these techniques be utilized, how-
ever, the application environment will dictate suitability. For
protection against sudden on-card load dumps at the DRAIN
pin of the MIC2588/MIC2594 controller, a 68V, 1W, 5% Zener
diode clamp (D2) connected from the DRAIN to the VEE of
the controller can be implemented, as shown. To protect
the controller from large-scale transients at the card input,
a 100V clamp diode (D1, SMAT70A or equivalent) can be
used. In either case, very short lead lengths and compact
layout design is strongly recommended to prevent unwanted
transients in the protection circuitry. Power buss inductance
often produces localized (plug-in card) high-voltage transients
during a turn-off event. Managing these repeated voltage
stresses with sufficient input bulk capacitance and/or tran-
sient suppressing diode clamps is highly recommended for
maximizing the life of the hot swap controller(s).
/PWRGD
DRAIN
UV
OV
VDD
SENSE
GATE
VEE
1 8
7
6
54
3
2
R1
698kΩ
1%
R3
12.4kΩ
1%
R2
11.8kΩ
1%
C1
0.47µF
*C6
0.47uF
*R5
47kΩ
*D1
100V
M1
SUM110N10-09
C3
0.33uF
R4
10Ω
-48VOUT
-48V RTN
C5
47uF C4
0.1uF
-48VIN
-48V RTN
C2
0.22uF
MIC2588-2BM
*R6
2.7kΩ
System
Reset
RSENSE
0.01Ω
5%
*M2 RFDBK
10kΩ
CFDBK
10nF
100V
*R7
10kΩ
* Optional components (See Applications Information for more details)
An SOT-363 is recommended for M2.
D2 is a 68V, 1W Zener diode.D2 is
*D2
68V
Figure 6 Optional Components for Added Performance/Protection
September 2005 15
September 2005 15 M9999-083005
MIC2588/MIC2594 Micrel
For systems that experience known load current surges
exceeding the 400µs internal overcurrent filter (tFLT), the
RC circuit consisting of R6 and C6 provides a means for
additional overcurrent filtering to eliminate false “tripping”
of the circuit breaker due to these transient load current
surges. It is highly recommended to limit the increase of
the overcurrent filter to approximately 2x the internal filter to
allow the MOSFET to operate within its thermal specifica-
tions and SOA. R6 and C6 act as a low-pass filter to reduce
the slew rate of the SENSE pin voltage. The SENSE pin
current is nominally 200nA, resulting in a slight voltage drop
across R6 that will combine in series with the voltage across
RSENSE to produce an effective circuit breaker trip voltage of
VTRIP – (R6 × ISENSE). The following equation can be used
to select component values for a given overcurrent filter
delay.
– (R × C) ln 1 – VTRIP – V(tO)
V(t) – V(tO)
tOCDLY
(8)
where VTRIP is the typical circuit breaker trip voltage specified
in the electrical specifications, V(t0) is the voltage drop across
the sense resister before the short or overcurrent condition
occurs, and V(t) is the voltage drop across the sense resis-
tor when the short or overcurrent is applied. The following
example sets an overcurrent delay of 1ms for a 7.5A load
current surge with a 2A steady-state load current and 5A
current limit (RSENSE = 10mΩ).
VTRIP = 50mV
V(t0) = 2A × 10mΩ = 20mV
V(t) = 7.5A × 10mΩ = 75mV
Using Equation 8, for R6 = 2.7kΩ, C6 is 0.47µF.
The capacitor (C2) connected from UV to reference (VEE)
is used as a glitch filter for the input undervoltage monitor.
C2 combines with the resistive network at the UV pin to
form an RC time constant to slow the UV pin voltage fall
time whenever the input voltage experiences a negative
(magnitude) transient. During start-up, the UV rise time will
also be affected by a longer RC time constant due to R1,
therefore, the output start cycle will be delayed until the UV
pin crosses its threshold.
The circuit in Figure 6 consisting of M2, R7, R8, and a digital
control signal, can be used to reset the controller after the
GATE (and output) turns off. Once the output has been
latched off, applying a low-high-low pulse on the GATE of M2
via the System Enable control can toggle the UV pin. System
Enable is a user defined signal referenced to VEE.
Sense Resistor Selection
The sense resistor is nominally valued at:
VTRIP(typ)
IHOT_SWAP(nom)
RSENSE(nom)
(9)
where VTRIP(TYP) is the typical (or nominal) circuit breaker
threshold voltage (50mV) and IHOT_SWAP(NOM) is the nomi-
nal load current level necessary to trip the internal circuit
breaker.
To accommodate worse-case tolerances in the sense re-
sistor (for a ±1% initial tolerance, allow ±3% tolerance for
variations over time and temperature) and circuit breaker
threshold voltages, a slightly more detailed calculation must
be used to determine the minimum and maximum hot swap
load currents.
As the MIC2588’s minimum current limit threshold voltage
is 40mV, the minimum hot swap load current is determined
where the sense resistor is 3% high:
40mV
1.03 × RSENSE(nom)
38.8mV
RSENSE(nom)
IHOT_SWAP(min)
 
Keep in mind that the minimum hot swap load current
should be greater than the application circuit’s upper
steady-state load current boundary. Once the lower value
of RSENSE has been calculated, it is good practice to check
the maximum hot swap load current (IHOT_SWAP(MAX)) that
the circuit may let pass in the case of tolerance build-up in
the opposite direction. Here, the worse case maximum is
found using a VTRIP(MAX) threshold of 60mV and a sense
resistor 3% low in value:
60mV
0.97 × RSENSE(nom)
61.9mV
RSENSE(nom)
IHOT_SWAP(max)
 
In this case, the application circuit must be sturdy enough to
operate up to approximately 1.5x the steady-state hot swap
load current. For example, if an MIC2588 circuit must pass a
minimum hot swap load current of 4A without nuisance trips,
RSENSE should be set to:
40mV
4A 10mΩ
RSENSE(nom)  
where the nearest 1% standard value is 10.0mΩ. At the
other tolerance extremes, IHOT_SWAP(MAX) for the circuit in
question is then simply:
61.9mV
10mΩ 6.19A
IHOT_SWAP(max)  
With a knowledge of the application circuit’s maximum
hot swap load current, the power dissipation rating of the
sense resistor can be determined using P = I2 × R. Here,
the current is IHOT_SWAP(max) = 6.19A and the resistance
RSENSE(max) = (1.03)(RSENSE(nom)) = 10.3mΩ.
Thus, the sense resistor’s maximum power dissipation is:
PMAX = (6.19A)2 × (10.3mΩ) = 0.395W
A 0.5W sense resistor is a good choice in this application.
Power MOSFET Selection
Selecting the proper external MOSFET for use with the-
MIC2588/MIC2594 involves three straightforward tasks:
•Choice of a MOSFET which meets minimum voltage
requirements.
•Selection of a device to handle the maximum continuous
current (steady-state thermal issues).
•Verify the selected part’s ability to withstand any peak
currents (transient thermal issues).
Power MOSFET Operating Voltage Requirements
The first voltage requirement for the MOSFET is easily stated:
September 2005 16
September 2005 16 M9999-083005
MIC2588/MIC2594 Micrel
the drain-source breakdown voltage of the MOSFET must
be greater than VIN(MAX), or VDD – VEE(min).
The second breakdown voltage criterion that must be met
is the gate-source voltage. For the MIC2588/MIC2594, the
gate of the external MOSFET is driven up to a maximum of
11V above VEE. This means that the external MOSFET must
be chosen to have a gate-source breakdown voltage of 12V
or more; 20V is recommended. Most power MOSFETs with
a 20V gate-source voltage rating have a 30V drain-source
breakdown rating or higher. For many 48V telecom applica-
tions, transient voltage spikes can approach, and sometimes
exceed, 100V. The absolute maximum input voltage rating
of the MIC2588/MIC2594 is 100V; therefore, a drain-source
breakdown voltage of 100V is suggested for the external
MOSFET. Additionally, an external input voltage clamp is
strongly recommended for applications that do not utilize
conditioned power supplies.
Power MOSFET Steady-State Thermal Issues
The selection of a MOSFET to meet the maximum continuous
current is a fairly straightforward exercise. First, arm yourself
with the following data:
•The value of ILOAD(CONT, MAX.) for the output in question
(see Sense Resistor Selection).
•The manufacturer’s datasheet for the candidate MOS-
FET.
•The maximum ambient temperature in which the device
will be required to operate.
•Any knowledge you can get about the heat sinking avail-
able to the device (e.g., can heat be dissipated into the
ground plane or power plane, if using a surface-mount
part? Is any airflow available?).
The datasheet will almost always give a value of on resistance
for a given MOSFET at a gate-source voltage of 4.5V and
10V. For MIC2588/MIC2594 applications, choose the gate-
source ON resistance at 10V and call this value RON. Since
a heavily enhanced MOSFET acts as an ohmic (resistive)
device, almost all that’s required to determine steady-state
power dissipation is to calculate I2R. The one addendum to
this is that MOSFETs have a slight increase in RON with in-
creasing die temperature. A good approximation for this value
is 0.5% increase in RON per °C rise in junction temperature
above the point at which RON was initially specified by the
manufacturer. For instance, if the selected MOSFET has a
calculated RON of 10mΩ at aTJ = 25°C, and the actual junc-
tion temperature ends up at 110°C, a good first cut at the
operating value for RON would be:
RON ≈ 10mΩ[1 + (110 – 25)(0.005)] ≈14.3mΩ
The final step is to make sure that the heat sinking available
to the MOSFET is capable of dissipating at least as much
power (rated in °C/W) as that with which the MOSFET’s
performance was specified by the manufacturer. Here are
a few practical tips:
1. The heat from a TO-263 power MOSFET flows
almost entirely out of the drain tab. If the drain tab
can be soldered down to one square inch or more,
the copper will act as the heat sink for the part. This
copper must be on the same layer of the board as
the MOSFET drain.
2. Airflow works. Even a few LFM (linear feet per-
minute) of air will cool a MOSFET down substan-
tially. If you can, position the MOSFET(s) near
the inlet of a power supply’s fan, or the outlet of a
processor’s cooling fan.
3. The best test of a candidate MOSFET for an ap-
plication (assuming the above tips show it to be a
likely fit) is an empirical one. Check the MOSFET’s
temperature in the actual layout of the expected
final circuit, at full operating current. The use of a
thermocouple on the drain leads, or infrared pyrom-
eter on the package, will then give a reasonable
idea of the device’s junction temperature.
Power MOSFET Transient Thermal Issues
If the prospecitve MOSFET has been shown to withstand the
environmental voltage stresses and the worse-case steady-
state power dissipation is addressed, the remaining task is to
verify if the MOSFET is capable of handling extreme overcur-
rent load faults, such as a short circuit, without overheating.
A power MOSFET can handle a much higher pulsed power
without damage than its continuos power dissipation ratings
imply due to an inherent trait, thermal inertia. With respect to
the specification and use of power MOSFETs, the parameter
of interest is the “Transient Thermal Impedence”, or Zθ, which
is a real number (variable factor) used as a multiplier of the
thermal resistance (Rθ). The multiplier is determined using
the given “Transient Thermal Imepedence Graph”, normalized
to Rθ, that displays curves for the thermal impedence versus
power pulse duration and duty cycle. The single-pulse curve
is appropriate for most hot swap applications. Zθ is specified
from junction-to-case for power MOSFETs typically used in
telecom applications.
The following example provides a method for estimating the
peak junction temperature of a power MOSFET in determin-
ing if the MOSFET is suitable for a particular application.
VIN (VDD VEE) = 48V, ILIM = 4.2A, and the power MOSFET
is SUM110N10-09 (TO-263 package) from Vishay-Siliconix.
This MOSFET has an RON of 9.5mΩ (TJ = 25°C), the junc-
tion-to-case thermal resistance (Rθ(J-C)) is 0.4°C/W, junc-
tion-to-ambient thermal resistance (Rθ(J-A)) is 40°C/W, and
the Transient Thermal Impedence Curve is shown in Figure
7. Consider, say, the MOSFET is switched on at time t1 and
the steady-state load current passing through the MOSFET
is 3A. At some point in time after t1, at time t2, there is an
unexpected short-circuit applied to the load, causing the
MIC2588/MIC2594 controller to adjust the GATE output
voltage and regulate the load current for 400µs at the pro-
grammed current limit value, 4.2A in this example. During this
short-circuit load condition, the dissipation in the MOSFET
is calculated by:
PD(short) = VDS × ILIM VDS = 0V – (-48V) = 48V
PD(short) = 48V × 4.2A = 201.6W for 400µs.
At first glance, it would appear that a very hefty MOSFET is
required to withstand this extreme overload condition. Upon
further examination, the calculation to approximate the peak
junction temperature is not a difficult task. The first step is to
determine the maximum steady-state junction temperature,
2 Km MIWMIWN‘W ‘ 0w apnea n: n' mun-mg: [Ir-cw- yum-m r urn-A hum EJ" 1:4 aua:m=uoemasmxm September 2005 17
September 2005 17 M9999-083005
MIC2588/MIC2594 Micrel
then add the rise in temperature due to the maximum power
dissipated during a transient overload caused by a short
circuit condtion. The equation to estimate the maximum
steady-state junction temperature is given by:
TJ(steady-state) TC(max) + ΔTJ (10)
TC(max) is the highest anticipated case temperaure, prior to
an overcurrent condition, at which the MOSFET will operate
and is estimated from the following equation based on the
highest ambient temperature of the system environment.
TC(max) = TA(max) + PD × (Rθ(J-A) – Rθ(J-C)) (11)
Let’s assume a maximum ambient of 60°C. The power dis-
sipation of the MOSFET is determined by the current through
the MOSFET and the on-resistance (I2R), which we will esti-
mate at 17mΩ (specification given at TJ = 125°C). Using our
example information and substituting into Equation 11,
TC(max) = 60°C + [((3A)2 × 17mΩ) × (40 0.4)°C/W]
= 66.06°C
Substituting the variables into Equation 10, TJ is determined
by:
TJ(steady-state) TC(max)+[RON+(TC(max)–TC)(0.005)
× (RON)][I2×(Rθ(J-A)–Rθ(J-C))]
66.06°C+[17mΩ+(66.06°C–25°C)(0.005/°C)
× (17mΩ)][(3A)2×(40–0.4)°C/W]
66.06°C + 7.30°C
73.36°C
Since this is not a closed-form equation, getting a close ap-
poroximation may take one or two iterations. On the second
iteration, start with TJ equal to the value calculated above.
Doing so in this example yields;
TJ(steady-state) 66.06°C+[17mΩ+(73.36°C–25°C)×(0.005/°C)
×(17mΩ)][(3A)2×(40–0.4)]°C/W
73.62°C
Another iteration shows that the result (73.63°C) is converg-
ing quickly, so we’ll estimate the maximum TJ(steady-state) at
74°C.
The use of the Transient Thermal Impedence Curves is
necessary to determine the increase in junction temperature
associated with a worst-case transient condition. From our
previous calculation of the maximum power dissipated during
a short circuit event for the MIC2588/MIC2594, we calculate
the transient junction temperature increase as:
TJ(transient) = PD(short) × Rθ(J-C) × Multiplier (12)
Assume the MOSFET has been on for a long time several
minutes or more and delivering the steady-state load current
of 3A to the load when the load is short circuited. The control-
ler will regulate the GATE output voltage to limit the current
to the programmed value of 4.2A for approximately 400µs
before immediately shutting off the output. For this situation
and almost all hot swap applications, this can be considered a
single pulse event as there is no significant duty cycle. From
Figure 7, find the point on the X-axis (“Square-Wave Pulse
Duration”) for 1ms, allowing for a healthy margin of the 400µs
tFLT, and read up the Y-axis scale to find the intersection of
the Single Pulse curve. This point is the normalized transient
thermal impedence (Zθ(J-C)), and the effective transient thermal
impedence is the product of Rθ(J-C) and the multiplier, 0.45
in this example. Solving Equation 12,
TJ(transient) = (201.6W) × (0.4°C/W) × 0.45 = 36.3°C
Finally, add this result to the maximum steady state junction
temperature calculated previously to determine the estimated
maximum transient junction temperature of the MOSFET:
TJ(max.transient) = 74°C + 36.3°C = 110.3°C, which is safely
under the specified maximum junction temperature of 200°C
for the SUM110N10-09.
FIgure 7. Transient Thermal Impedance - SUM110N10-09
Cunem Haw «a me Lead jjjj Cmvem me «am we Luau 18 September 2005
September 2005 18 M9999-083005
MIC2588/MIC2594 Micrel
PCB Layout Considerations
4-Wire Kelvin Sensing
Because of the low value typically required for the sense
resistor, special care must be used to measure accurately
the voltage drop across it. Specifically, the measurement
technique across each RSENSE must employ 4-wire Kelvin
sensing. This is simply a means of making sure that any
voltage drops in the power traces connecting to the resistors
are not picked up by the signal conductors measuring the
voltages across the sense resistors.
Figure 8 illustrates how to implement 4-wire Kelvin sensing.
As the figure shows, all the high current in the circuit (from
VEE through RSENSE, and then to the source of the output
MOSFET) flows directly through the power PCB traces
and RSENSE. The voltage drop resulting across RSENSE is
sampled in such a way that the high currents through the
power traces will not introduce any parasitic voltage drops
in the sense leads. It is recommended to connect the hot
swap controller’s sense leads directly to the sense resistor’s
metalized contact pads.
Other Layout Considerations
Figure 9 is a suggested PCB layout diagram for the MIC2588/
MIC2594. Many hot swap applications will require load currents
of several amperes. Therefore, the power (VEE and Return)
trace widths (W) need to be wide enough to allow the current
to flow while the rise in temperature for a given copper plate
(e.g., 1oz. or 2oz.) is kept to a maximum of 10°C to 25°C.
The return (or power ground) trace should be the same width
as the positive voltage power traces (input/load) and isolated
from any ground and signal planes so that the controller’s
power is common mode. Also, these traces should be as
short as possible in order to minimize the IR drops between
the input and the load.
Finally, the use of plated-through vias will be necessary to
make circuit connections to the power, ground and signal
planes of multi-layer PCBs.
RSENSE
Power Trace
From VEE
PCB Track Width:
0.03" per Ampere
using 1oz Cu
Power Trace
To MOSFET Source
Signal Trace
to MIC2588/MIC2594 VEE Pin Signal Trace
to MIC2588/MIC2594 SENSE Pin
Note: Each SENSE lead trace shall be
balanced for best performance with equal
length/equal aspect ratio.
R
SENSE
metalized
contact pads
Figure 8. 4-Wire Kelvin Sense Connections for RSENSE
CFDBK
R4
*POWER MOSFET
(TO-263)
W
W
W
Via to the
power (VEE output)
plane
Current Flow
from the Load
Current Flow
to the Load
-DRAWING IS NOT TO SCALE-
*See Table 1 for part numbers and vendors
^R1 placed on bottom side
Power Plane -------- (red)
Ground Plane ------- (black)
Trace width (W) guidelines and additional information given in
"PCB Layout Recommendations" section of the datasheet
Via to the
bottom side
Via to the
ground plane
*SENSE RESISTOR
(WSR-2 or
WSL2512)
D1
MIC2588-2BM
VDD
DRAIN
GATE
OV
VEE
/PWRGD
UV
SENSE
RFDBK
R3 R2
^R1
C3
Via to the
Return (VDD)
plane
Via to the
power (VEE output)
plane
GROUND
PAD
C1
Via to the
Return (VDD)
plane
Figure 9. Recommended PCB Layout for Sense Resistor, Power MOSFET, Overvoltage/Undervoltage Resistive
Divider Network, and Timer Capacitors
September 2005 19
September 2005 19 M9999-083005
MIC2588/MIC2594 Micrel
MOSFET and Sense Resistor Vendors
Device types, part numbers, and manufacturer contacts
for power MOSFTETS and sense resistors are provided in
Table 1.
MOSFET Vendors Key MOSFET Type(s) Breakdown Voltage (VDSS) Contact Information
Vishay - Siliconix
SUM75N06-09L (TO-263)
SUM70N06-11 (TO-263)
SUM50N06-16L (TO-263)
60V
60V
60V
www.siliconix.com
(203) 452-5664
SUP85N10-10 (TO-220AB)
SUB85N10-10 (TO-263)
SUM110N10-09 (TO-263)
SUM60N10-17 (TO-263)
100V
100V
100V
100V
www.siliconix.com
(203) 452-5664
International Rectifier IRF530 (TO-220AB)
IRF540N (TO-220AB) 100V
100V www.irf.com
(310) 322-3331
Renesas 2SK1298 (TO-3PFM)
2SK1302 (TO-220AB)
2SK1304 (TO-3P)
60V
100V
100V
www.renesas.com
(408) 433-1990
Resistor Vendors Sense Resistors Contact Information
Vishay - Dale “WSL” and “WSR”
Series www.vishay.com/docswsl_30100.pdf
(203) 452-5664
IRC “OARS” Series
“LR” Series
second source to “WSL”
www.irctt.com/pdf_files/OARS.pdf
www.irctt.com/pdf_files/LRC.pdf
(828) 264-8861
Table 1. MOSFET and Sense Resistor Vendors
September 2005 20
September 2005 20 M9999-083005
MIC2588/MIC2594 Micrel
/PWRGD Signal Drive Capability
The /PWRGD signal can be used to drive an optoisolator or
an LED. The use of an optoisolator is sometimes needed
to protect I/O signals (e.g., /PWRGD, RESET, ENABLE) of
both the controller and downstream DC-DC converter(s)
from damage caused by common mode transients. Such
/PWRGD
DRAIN
UV
OV
VDD
SENSE
GATE
VEE
1 8
7
6
54
3
2
R1
698kΩ
1%
R3
12.4kΩ
1%
R2
11.8 kΩ
1%
C1
1uF
RSENSE
0.01Ω
5%
*D1
SMAT70A
100V
M1
SUM110N10-09
RFDBK
15kΩ
C3
0.22uF
CFDBK
6.8nF
100V
R4
10Ω
-48VOUT
-48V RTN
C5
100uFC4
0.1uF
-48VIN
(Long Pin)
-48V RTN
(Long Pin)
*C6
0.33uF
MIC2588-2BM
-48V RTN
(Short Pin)
DC-DC
CONVERTER
IN-
IN+
ON/OFF#
OUT+
OUT-
+2.5V RTN
+2.5VOUT
Nominal Undervoltage and Overvoltage Thresholds:
VUV = 36.5V
VOV = 71.2V
* Optional components (See Applications Information for more details)
# An external pull-up resistor for the power-good signal is necessary for DC-DC supplies
(and all other load modules) not equipped with internal pull-up impedence
VDD
*C2
22nF
1 6
52
MOC207-M
R5
43kΩ
Figure 10. Optoisolator Driven by /PWRGD Signal
is the case when an EMI filter is utilized to prevent DC-DC
converter switching noise from being injected back onto the
power supply. The circuit of Figure 10 shows how to con-
figure an optoisolator driven by the /PWRGD signal of the
MIC2588 controller.
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MIC2588/MIC2594 Micrel
M9999-083005 21 September 2005
Package Information
45
0–8
0.244 (6.20)
0.228 (5.79)
0.197 (5.0)
0.189 (4.8) SEATING
PLANE
0.026 (0.65)
MAX)
0.010 (0.25)
0.007 (0.18)
0.064 (1.63)
0.045 (1.14)
0.0098 (0.249)
0.0040 (0.102)
0.020 (0.51)
0.013 (0.33)
0.157 (3.99)
0.150 (3.81)
0.050 (1.27)
TYP
PIN 1
DIMENSIONS:
INCHES (MM)
0.050 (1.27)
0.016 (0.40)
8-Pin SOIC (M)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL + 1 (408) 944-0800 FAX + 1 (408) 474-1000 WEB http://www.micrel.com
The information furnished by Micrel in this datasheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its use.
Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.
Micrel Products are not
reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into
the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser’s
use or sale of Micrel Products for use in life support appliances, devices or systems is at Purchaser’s own risk and Purchaser agrees to fully indemnify
Micrel for any damages resulting from such use or sale.
© 2005 Micrel, Incorporated.

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