LT8300 Datasheet by Analog Devices Inc.

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ANALOG L POWER EY DEVICES ‘ D LINEAR“ LT8300 LAW
LT8300
1
Rev. A
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TYPICAL APPLICATION
FEATURES DESCRIPTION
100VIN Micropower Isolated Flyback
Converter with 150V/260mA Switch
The LT
®
8300 is a micropower high voltage isolated fly-
back converter. By sampling the isolated output voltage
directly from the primary-side flyback waveform, the part
requires no third winding or opto-isolator for regulation.
The output voltage is programmed with a single exter-
nal resistor. Internal compensation and soft-start further
reduce external component count. Boundary mode opera-
tion provides a small magnetic solution with excellent load
regulation. Low ripple Burst Mode operation maintains
high efficiency at light load while minimizing the output
voltage ripple. A 260mA, 150V DMOS power switch is
integrated along with all high voltage circuitry and control
logic into a 5-lead ThinSOT™ package.
The LT8300 operates from an input voltages range of 6V
to 100V and can deliver up to 2W of isolated output power.
The high level of integration and the use of boundary
and low ripple burst modes result in a simple to use, low
component count, and high efficiency application solution
for isolated power delivery.
5V Micropower Isolated Flyback Converter
APPLICATIONS
n 6V to 100V Input Voltage Range
n 260mA, 150V Internal DMOS Power Switch
n Low Quiescent Current
n 70µA in Sleep Mode
n 330µA in Active Mode
n Boundary Mode Operation at Heavy Load
n Low-Ripple Burst Mode
®
Operation at Light Load
n Minimum Load <0.5% (Typ) of Full Output
n VOUT Set with a Single External Resistor
n No Transformer Third Winding or Opto-Isolator
Required for Regulation
n Accurate EN/UVLO Threshold and Hysteresis
n Internal Compensation and Soft-Start
n 5-Lead TSOT-23 Package
n AEC-Q100 Qualified for Automotive Applications
n Isolated Telecom, Automotive, Industrial, Medical
Power Supplies
n Isolated Auxiliary/Housekeeping Power Supplies
Efficiency vs Load Current
LT8300
4:1
RFB
SW
300µH 19µH
EN/UVLO
1M
2.2µF
40.2k
VIN
VIN
36V TO 72V
VOUT
+
5V
1mA TO 300mA
VOUT
GND
210k
47µF
8300 TA01a
LOAD CURRENT (mA)
EFFICIENCY (%)
100
20
30
90
40
10
60
70
80
50
0
8300 TA01b
300
250200150100500
VIN = 48V
VIN = 72V
VIN = 36V
All registered trademarks and trademarks are the property of their respective owners. Protected
by U.S. Patents, including 5438499, 7463497, and 7471522.
LT8300 mp quw :I WU :I
LT8300
2
Rev. A
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PIN CONFIGURATIONABSOLUTE MAXIMUM RATINGS
SW (Note 2) ........................................................... 150V
VIN ......................................................................... 100V
EN/UVLO ................................................................... VIN
RFB ...................................................... VIN0.5V to VIN
Current into RFB ................................................... 200µA
Operating Junction Temperature Range (Notes 3, 4)
LT8300E, LT8300I ............................. 40°C to 125°C
LT8300H ............................................ 40°C to 150°C
LT8300MP ......................................... 55°C to 150°C
Storage Temperature Range .................. 65°C to 150°C
(Note 1)
EN/UVLO 1
GND 2
TOP VIEW
S5 PACKAGE
5-LEAD PLASTIC TSOT-23
TJMAX = 150°C, θJA
= 150°C/W
RFB 3
5 VIN
4 SW
ORDER INFORMATION
LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE
LT8300ES5#PBF LT8300ES5#TRPBF LTGFF 5-Lead Plastic TSOT-23 –40°C to 125°C
LT8300IS5#PBF LT8300IS5#TRPBF LTGFF 5-Lead Plastic TSOT-23 –40°C to 125°C
LT8300HS5#PBF LT8300HS5#TRPBF LTGFF 5-Lead Plastic TSOT-23 –40°C to 150°C
LT8300MPS5#PBF LT8300MPS5#TRPBF LTGFF 5-Lead Plastic TSOT-23 –55°C to 150°C
AUTOMOTIVE PRODUCTS**
LEAD FREE FINISH TAPE AND REEL TAPE AND REEL MINI PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE
LT8300ES5#WPBF LT8300ES5#WTRPBF LT8300ES5#WTRMPBF LTGFF 5-Lead Plastic TSOT-23 –40°C to 125°C
LT8300IS5#WPBF LT8300IS5#WTRPBF LT8300IS5#WTRMPBF LTGFF 5-Lead Plastic TSOT-23 –40°C to 125°C
LT8300HS5#WPBF LT8300HS5#WTRPBF LT8300HS5#WTRMPBF LTGFF 5-Lead Plastic TSOT-23 –40°C to 150°C
Contact the factory for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Tape and reel specifications. Some packages are available in 500 unit reels through designated sales channels with #TRMPBF suffix.
**Versions of this part are available with controlled manufacturing to support the quality and reliability requirements of automotive applications. These
models are designated with a #W suffix. Only the automotive grade products shown are available for use in automotive applications. Contact your
local Analog Devices account representative for specific product ordering information and to obtain the specific Automotive Reliability reports for
thesemodels.
LT8300
LT8300
3
Rev. A
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ELECTRICAL CHARACTERISTICS
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The SW pin is rated to 150V for transients. Depending on the
leakage inductance voltage spike, operating waveforms of the SW pin
should be derated to keep the flyback voltage spike below 150V as shown
in Figure5.
Note 3: The LT8300E is guaranteed to meet performance specifications
from 0°C to 125°C operating junction temperature. Specifications over
the –40°C to 125°C operating junction temperature range are assured by
design, characterization and correlation with statistical process controls.
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 24V, VEN/UVLO = VIN unless otherwise noted.
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNIT
VIN Input Voltage Range 6 100 V
VIN UVLO Threshold Rising
Falling
5.8
3.2
6 V
V
IQVIN Quiescent Current VEN/UVLO = 0.3V
VEN/UVLO = 1.1V
Sleep Mode (Switch Off)
Active Mode (Switch On)
1.2
200
70
330
2 µA
µA
µA
µA
EN/UVLO Shutdown Threshold For Lowest Off IQl0.3 0.75 V
EN/UVLO Enable Threshold Falling
Hysteresis
l1.199 1.223
0.016
1.270 V
V
IHYS EN/UVLO Hysteresis Current VEN/UVLO = 0.3V
VEN/UVLO = 1.1V
VEN/UVLO = 1.3V
–0.1
2.2
–0.1
0
2.5
0
0.1
2.8
0.1
µA
µA
µA
fMAX Maximum Switching Frequency 720 750 780 kHz
fMIN Minimum Switching Frequency 6 7.5 9 kHz
tON(MIN) Minimum Switch-On Time 160 ns
tOFF(MIN) Minimum Switch-Off Time 350 ns
tOFF(MAX) Maximum Switch-Off Time Backup Timer 200 µs
ISW(MAX) Maximum SW Current Limit l228 260 292 mA
ISW(MIN) Minimum SW Current Limit l34 52 70 mA
SW Over Current Limit To Initiate Soft-Start 520 mA
RDS(ON) Switch On-Resistance ISW = 100mA 10 Ω
ILKG Switch Leakage Current VIN = 100V, VSW = 150V 0.1 0.5 µA
IRFB RFB Regulation Current l98 100 102 µA
RFB Regulation Current Line Regulation 6V ≤ VIN ≤ 100V 0.001 0.01 %/V
tSS Soft-Start Timer 2.7 ms
The LT8300I is guaranteed over the full –40°C to 125°C operating junction
temperature range. The LT8300H is guaranteed over the full –40°C to
150°C operating junction temperature range. The LT8300MP is guaranteed
over the full –55°C to 150°C operating junction temperature range. High
junction temperatures degrade operating lifetimes. Operating lifetime is
derated at junction temperature greater than 125°C.
Note 4: The LT8300 includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed 150°C when overtemperature protection is active.
Continuous operation above the specified maximum operating junction
temperature may impair device reliability.
LT8300 / 500 520 3% $5255 S WEE? SE3 S #25? 51:6
LT8300
4
Rev. A
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TYPICAL PERFORMANCE CHARACTERISTICS
Boundary Mode Waveforms Discontinuous Mode Waveforms Burst Mode Waveforms
VIN Shutdown Current
VIN Quiescent Current,
Sleep Mode
VIN Quiescent Current,
Active Mode
Output Load and Line Regulation Output Temperature Variation
Switching Frequency
vs Load Current
TA = 25°C, unless otherwise noted.
LOAD CURRENT (mA)
OUTPUT VOLTAGE (V)
5.20
4.85
5.15
4.90
5.00
5.05
5.10
4.95
4.80
8300 G01
300
250200150100500
VIN = 36V
VIN = 48V
VIN = 72V
FRONT PAGE APPLICATION
AMBIENT TEMPERATURE (°C)
OUTPUT VOLTAGE (V)
5.5
4.6
5.4
4.7
5.0
5.1
5.2
5.3
4.8
4.9
4.5
8300 G02
150
7550 125100250–25–50
FRONT PAGE APPLICATION
VIN = 48V, IOUT = 200mA
LOAD CURRENT (mA)
FREQUENCY (kHz)
500
100
200
400
300
0
8300 G03
300
250200150100500
FRONT PAGE APPLICATION
VIN = 48V
2µs/DIV
FRONT PAGE APPLICATION
VIN = 48V, IOUT = 300mA
ILPRI
100mA/DIV
VSW
50V/DIV
VOUT
50mV/DIV
8300 G04 2µs/DIV
FRONT PAGE APPLICATION
VIN = 48V, IOUT = 60mA
ILPRI
100mA/DIV
VSW
50V/DIV
VOUT
50mV/DIV
8300 G05
20µs/DIV
FRONT PAGE APPLICATION
VIN = 48V, IOUT = 1mA
ILPRI
100mA/DIV
VSW
50V/DIV
VOUT
50mV/DIV
8300 G06
VIN (V)
IQ (µA)
10
2
4
8
6
0
8300 G07
100806040200
TJ = –55°C
TJ = 150°C TJ = 25°C
VIN (V)
IQ (µA)
100
50
60
80
90
70
40
8300 G08
100806040200
TJ = 25°C
TJ = –55°C
TJ = 150°C
VIN (V)
IQ (µA)
380
300
320
360
340
280
100806040200
TJ = 25°C
TJ = –55°C
TJ = 150°C
LT8300 ms 23 m _ 23 i _ \ 2m man :15 555$: 300 .552, _ 25 .9 5.25m? man :5 m2: 400 :5 WE: 20 33 553E
LT8300
5
Rev. A
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TYPICAL PERFORMANCE CHARACTERISTICS
RDS(ON) Switch Current Limit Maximum Switching Frequency
Minimum Switching Frequency Minimum Switch-On Time Minimum Switch-Off Time
EN/UVLO Enable Threshold EN/UVLO Hysteresis Current RFB Regulation Current
TA = 25°C, unless otherwise noted.
TEMPERATURE (°C)
V
EN/UVLO
(V)
1.240
1.205
1.210
1.225
1.230
1.235
1.215
1.220
1.200
8300 G10
150
7550 125100250–25–50
TEMPERATURE (°C)
I
HYS
(µA)
5
1
2
3
4
0
8300 G11
150
7550 125100250–25–50
TEMPERATURE (°C)
I
RFB
(µA)
105
101
102
103
104
100
96
97
98
99
95
8300 G12
150
7550 125100250–25–50
TEMPERATURE (°C)
RESISTANCE (Ω)
25
5
10
15
20
0
8300 G13
150
7550 125100250–25–50
TEMPERATURE (°C)
I
SW
(mA)
300
50
100
150
200
250
0
8300 G14
150
7550 125100250–25–50
MAXIMUM CURRENT LIMIT
MINIMUM CURRENT LIMIT
TEMPERATURE (°C)
FREQUENCY (kHz)
1000
200
400
600
800
0
8300 G15
150
7550 125100250–25–50
TEMPERATURE (°C)
FREQUENCY (kHz)
20
4
8
12
16
0
8300 G16
150
7550 125100250–25–50
TEMPERATURE (°C)
TIME (ns)
400
100
200
300
0
8300 G17
150
7550 125100250–25–50
TEMPERATURE (°C)
TIME (ns)
400
100
200
300
0
8300 G18
150
7550 125100250–25–50
em 1. I.
LT8300
6
Rev. A
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PIN FUNCTIONS
EN/UVLO (Pin 1): Enable/Undervoltage Lockout. The
EN/UVLO pin is used to enable the LT8300. Pull the pin
below 0.3V to shut down the LT8300. This pin has an
accurate 1.223V threshold and can be used to program a
VIN undervoltage lockout (UVLO) threshold using a resis-
tor divider from VIN to ground. A 2.5µA current hysteresis
allows the programming of V
IN
UVLO hysteresis. If neither
function is used, tie this pin directly to VIN.
GND (Pin 2): Ground. Tie this pin directly to local ground
plane.
RFB (Pin 3): Input Pin for External Feedback Resistor.
Connect a resistor from this pin to the transformer
primary SW pin. The ratio of the RFB resistor to the inter-
nal trimmed 12.23k resistor, times the internal bandgap
reference, determines the output voltage (plus the effect
of any non-unity transformer turns ratio). Minimize trace
area at this pin.
SW (Pin 4): Drain of the 150V Internal DMOS Power
Switch. Minimize trace area at this pin to reduce EMI and
voltage spikes.
V
IN
(Pin 5): Input Supply. The V
IN
pin supplies current
to internal circuitry and serves as a reference voltage for
the feedback circuitry connected to the RFB pin. Locally
bypass this pin to ground with a capacitor.
BLOCK DIAGRAM
8300 BD
+
+
OSCILLATOR
1:4
S
R Q
1.223V
25µA
M2M3
BOUNDARY
DETECTOR
DRIVER
+
A2
A3
RSENSE
0.3Ω
M1
gm
RREF
12.23kΩ
RFB
2.5µA
R2
EN/UVLO
M4
3 45
+
1.223V
A1
1
REFERENCE
REGULATORS
VIN
2
GND
RFB SWVIN
V
IN
T1
NPS:1
D
OUT
LSEC
LPRI
VOUT
+
VOUT
COUT
CIN
R1
LT8300
LT8300
7
Rev. A
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OPERATION
The LT8300 is a current mode switching regulator IC
designed specially for the isolated flyback topology. The
key problem in isolated topologies is how to communicate
the output voltage information from the isolated secondary
side of the transformer to the primary side for regulation.
Historically, opto-isolators or extra transformer windings
communicate this information across the isolation bound-
ary. Opto-isolator circuits waste output power, and the extra
components increase the cost and physical size of the
power supply. Opto-isolators can also cause system issues
due to limited dynamic response, nonlinearity, unit-to-unit
variation and aging over lifetime. Circuits employing extra
transformer windings also exhibit deficiencies, as using an
extra winding adds to the transformers physical size and
cost, and dynamic response is often mediocre.
The LT8300 samples the isolated output voltage through
the primary-side flyback pulse waveform. In this man-
ner, neither opto-isolator nor extra transformer winding
is required for regulation. Since the LT8300 operates
in either boundary conduction mode or discontinuous
conduction mode, the output voltage is always sampled
on the SW pin when the secondary current is zero. This
method improves load regulation without the need of
external load compensation components.
The LT8300 is a simple to use micropower isolated fly-
back converter housed in a 5-lead TSOT-23 package. The
output voltage is programmed with a single external resis-
tor. By integrating the loop compensation and soft-start
inside, the part further reduces the number of external
components. As shown in the Block Diagram, many of
the blocks are similar to those found in traditional switch-
ing regulators including reference, regulators, oscillator,
logic, current amplifier, current comparator, driver, and
power switch. The novel sections include a flyback pulse
sense circuit, a sample-and-hold error amplifier, and a
boundary mode detector, as well as the additional logic for
boundary conduction mode, discontinuous conduction
mode, and low ripple Burst Mode operation.
Boundary Conduction Mode Operation
The LT8300 features boundary conduction mode opera-
tion at heavy load, where the chip turns on the primary
power switch when the secondary current is zero.
Boundary conduction mode is a variable frequency, vari-
able peak-current switching scheme. The power switch
turns on and the transformer primary current increases
until an internally controlled peak current limit. After the
power switch turns off, the voltage on the SW pin rises to
the output voltage multiplied by the primary-to-secondary
transformer turns ratio plus the input voltage. When the
secondary current through the output diode falls to zero,
the SW pin voltage collapses and rings around VIN. A
boundary mode detector senses this event and turns the
power switch back on.
Boundary conduction mode returns the secondary current
to zero every cycle, so parasitic resistive voltage drops
do not cause load regulation errors. Boundary conduc-
tion mode also allows the use of smaller transformers
compared to continuous conduction mode and does not
exhibit subharmonic oscillation.
Discontinuous Conduction Mode Operation
As the load gets lighter, boundary conduction mode
increases the switching frequency and decreases the
switch peak current at the same ratio. Running at a higher
switching frequency up to several MHz increases switch-
ing and gate charge losses. To avoid this scenario, the
LT8300 has an additional internal oscillator, which clamps
the maximum switching frequency to be less than 750kHz.
Once the switching frequency hits the internal frequency
clamp, the part starts to delay the switch turn-on and
operates in discontinuous conduction mode.
Low Ripple Burst Mode Operation
Unlike traditional flyback converters, the LT8300 has to
turn on and off at least for a minimum amount of time
and with a minimum frequency to allow accurate sampling
of the output voltage. The inherent minimum switch cur-
rent limit and minimum switch-off time are necessary to
guarantee the correct operation of specific applications.
As the load gets very light, the LT8300 starts to fold
back the switching frequency while keeping the mini-
mum switch current limit. So the load current is able to
decrease while still allowing minimum switch-off time for
the sample-and-hold error amplifier. Meanwhile, the part
switches between sleep mode and active mode, thereby
LT8300 VFLBK VBG
LT8300
8
Rev. A
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OPERATION
reducing the effective quiescent current to improve light
load efficiency. In this condition, the LT8300 operates
in low ripple Burst Mode. The typical 7.5kHz minimum
Output Voltage
The RFB resistor as depicted in the Block Diagram is the
only external resistor used to program the output voltage.
The LT8300 operates similar to traditional current mode
switchers, except in the use of a unique flyback pulse
sense circuit and a sample-and-hold error amplifier, which
sample and therefore regulate the isolated output voltage
from the flyback pulse.
Operation is as follows: when the power switch M1 turns
off, the SW pin voltage rises above the VIN supply. The
amplitude of the flyback pulse, i.e., the difference between
the SW pin voltage and VIN supply, is given as:
VFLBK = (VOUT + VF + ISEC • ESR) • NPS
VF = Output diode forward voltage
ISEC = Transformer secondary current
ESR = Total impedance of secondary circuit
NPS = Transformer effective primary-to-secondary
turns ratio
The flyback voltage is then converted to a current IRFB by
the flyback pulse sense circuit (M2 and M3). This current
IRFB also flows through the internal trimmed 12.23k RREF
resistor to generate a ground-referred voltage. The result-
ing voltage feeds to the inverting input of the sample-
and-hold error amplifier. Since the sample-and-hold error
amplifier samples the voltage when the secondary current
is zero, the (I
SEC
ESR) term in the V
FLBK
equation can be
assumed to be zero.
The bandgap reference voltage VBG, 1.223V, feeds to the
non-inverting input of the sample-and-hold error ampli-
fier. The relatively high gain in the overall loop causes
the voltage across RREF resistor to be nearly equal to the
APPLICATIONS INFORMATION
switching frequency determines how often the output volt-
age is sampled and also the minimum load requirement.
bandgap reference voltage VBG. The resulting relationship
between VFLBK and VBG can be expressed as:
VFLBK
RFB
RREF =VBG
or
VFLBK =VBG
RREF
RFB =IRFB RFB
VBG = Bandgap reference voltage
IRFB = RFB regulation current = 100µA
Combination with the previous VFLBK equation yields an
equation for VOUT, in terms of the RFB resistor, trans-
former turns ratio, and diode forward voltage:
VOUT =10A • RFB
NPS
VF
Output Temperature Coefficient
The first term in the VOUT equation does not have tem-
perature dependence, but the output diode forward volt-
age VF has a significant negative temperature coefficient
(–1mVC to 2mVC). Such a negative temperature coef-
ficient produces approximately 200mV to 300mV voltage
variation on the output voltage across temperature.
For higher voltage outputs, such as 12V and 24V, the output
diode temperature coefficient has a negligible effect on the
output voltage regulation. For lower voltage outputs, such
as 3.3V and 5V, however, the output diode temperature
coefficient does count for an extra 2% to 5% output voltage
regulation. For customers requiring tight output voltage
regulation across temperature, please refer to other LTC
parts with integrated temperature compensation features.
LT8300 VOUT m
LT8300
9
Rev. A
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APPLICATIONS INFORMATION
Selecting Actual RFB Resistor Value
The LT8300 uses a unique sampling scheme to regulate
the isolated output voltage. Due to the sampling nature,
the scheme contains repeatable delays and error sources,
which will affect the output voltage and force a re-evalua-
tion of the RFB resistor value. Therefore, a simple two-step
process is required to choose feedback resistor RFB.
Rearrangement of the expression for VOUT in the Output
Voltage section yields the starting value for RFB:
RFB =NPS • VOUT +VF
( )
100µA
VOUT = Output voltage
VF = Output diode forward voltage = ~0.3V
NPS = Transformer effective primary-to-secondary
turns ratio
Power up the application with the starting RFB value and
other components connected, and measure the regulated
output voltage, VOUT(MEAS). The final RFB value can be
adjusted to:
RFB(FINAL) =VOUT
VOUT(MEAS)
RFB
Once the final RFB value is selected, the regulation accu-
racy from board to board for a given application will be
very consistent, typically under ±5% when including
device variation of all the components in the system
(assuming resistor tolerances and transformer windings
matching within ±1%). However, if the transformer or
the output diode is changed, or the layout is dramatically
altered, there may be some change in VOUT.
Output Power
A flyback converter has a complicated relationship
between the input and output currents compared to a
buck or a boost converter. A boost converter has a rela-
tively constant maximum input current regardless of input
voltage and a buck converter has a relatively constant
maximum output current regardless of input voltage. This
is due to the continuous non-switching behavior of the
two currents. A flyback converter has both discontinu-
ous input and output currents which make it similar to
a non-isolated buck-boost converter. The duty cycle will
affect the input and output currents, making it hard to
predict output power. In addition, the winding ratio can
be changed to multiply the output current at the expense
of a higher switch voltage.
The graphs in Figures 1 to 4 show the typical maximum
output power possible for the output voltages 3.3V, 5V,
12V, and 24V. The maximum output power curve is the
calculated output power if the switch voltage is 120V dur
-
ing the switch-off time. 30V of margin is left for leakage
inductance voltage spike. To achieve this power level at
a given input, a winding ratio value must be calculated
to stress the switch to 120V, resulting in some odd ratio
values. The curves below the maximum output power
curve are examples of common winding ratio values and
the amount of output power at given input voltages.
One design example would be a 5V output converter with
a minimum input voltage of 36V and a maximum input
voltage of 72V. A six-to-one winding ratio fits this design
example perfectly and outputs equal to 2.44W at 72V but
lowers to 1.87W at 36V.
The following equations calculate output power:
POUT = η • VIN D •ISW(MAX) • 0.5
η = Efficiency =85%
D=Duty Cycle =VOUT +VF
( )
NPS
VOUT +VF
( )
NPS +VIN
ISW(MAX) = Maximum switch current limit = 260mA
LT8300 33 ESE 5&3 4 / 7 1/ 35 :3 $38 5&8 :5 $251 5&8 :5 $251 5&8 “IO
LT8300
10
Rev. A
For more information www.analog.com
APPLICATIONS INFORMATION
Figure4. Output Power for 24V Output
Figure1. Output Power for 3.3V Output Figure2. Output Power for 5V Output
Figure3. Output Power for 12V Output
INPUT VOLTAGE (V)
0
OUTPUT POWER (W)
3.5
2.5
1.5
3.0
2.0
1.0
0.5
040 8020 60
8300 F01
100
N = 12:1
MAXIMUM
OUTPUT
POWER
N = 8:1
N = 6:1
N = 4:1
INPUT VOLTAGE (V)
0
OUTPUT POWER (W)
3.5
2.5
1.5
3.0
2.0
1.0
0.5
040 8020 60
8300 F02
100
MAXIMUM
OUTPUT
POWER
N = 2:1
N = 4:1
N = 6:1
N = 8:1
INPUT VOLTAGE (V)
0
OUTPUT POWER (W)
3.5
2.5
1.5
3.0
2.0
1.0
0.5
040 8020 60
8300 F03
100
N = 4:1
MAXIMUM
OUTPUT
POWER
N = 3:1
N = 2:1
N = 1:1
INPUT VOLTAGE (V)
0
OUTPUT POWER (W)
3.5
2.5
1.5
3.0
2.0
1.0
0.5
040 8020 60
8300 F04
100
N = 1:1
N = 3:2
N = 1:2
MAXIMUM
OUTPUT
POWER
N = 2:1
Primary Inductance Requirement
The LT8300 obtains output voltage information from the
reflected output voltage on the SW pin. The conduction of
secondary current reflects the output voltage on the pri-
mary SW pin. The sample-and-hold error amplifier needs
a minimum 350ns to settle and sample the reflected out-
put voltage. In order to ensure proper sampling, the sec-
ondary winding needs to conduct current for a minimum
of 350ns. The following equation gives the minimum
value for primary-side magnetizing inductance:
LPRI tOFF(MIN) NPS • VOUT +VF
( )
ISW(MIN)
tOFF(MIN) = Minimum switch-off time = 350ns
ISW(MIN) = Minimum switch current limit = 52mA
In addition to the primary inductance requirement for
the minimum switch-off time, the LT8300 has minimum
switch-on time that prevents the chip from turning on
the power switch shorter than approximately 160ns.
This minimum switch-on time is mainly for leading-edge
blanking the initial switch turn-on current spike. If the
inductor current exceeds the desired current limit during
that time, oscillation may occur at the output as the cur-
rent control loop will lose its ability to regulate. Therefore,
the following equation relating to maximum input voltage
must also be followed in selecting primary-side magnetiz-
ing inductance:
LPRI tON(MIN) • VIN(MAX)
ISW(MIN)
tON(MIN) = Minimum Switch-On Time = 160ns
LT8300 ‘I‘I
LT8300
11
Rev. A
For more information www.analog.com
APPLICATIONS INFORMATION
In general, choose a transformer with its primary mag-
netizing inductance about 20% to 40% larger than the
minimum values calculated above. A transformer with
much larger inductance will have a bigger physical size
and may cause instability at light load.
Selecting a Transformer
Transformer specification and design is perhaps the most
critical part of successfully applying the LT8300. In addi-
tion to the usual list of guidelines dealing with high fre-
quency isolated power supply transformer design, the
following information should be carefully considered.
Linear Technology has worked with several leading mag-
netic component manufacturers to produce pre-designed
flyback transformers for use with the LT8300. Table 1
shows the details of these transformers.
Turns Ratio
Note that when choosing the RFB resistor to set output
voltage, the user has relative freedom in selecting a trans-
former turns ratio to suit a given application. In contrast,
the use of simple ratios of small integers, e.g., 4:1, 2:1,
1:1, provides more freedom in settling total turns and
mutual inductance.
Table5. Predesigned Transformers — Typical Specifications
TRANSFORMER
PART NUMBER
LPRI
(µH)
LLEAKAGE
(µH) NP:NS:NB VENDOR TARGET APPLICATIONS
750312367 400 4.5 8:1 Wurth Elektronik 48V to 3.3V/0.51A, 24V to 3.3V/0.37A, 12V to 3.3V/0.24A
750312557 300 2.5 6:1 Wurth Elektronik 48V to 3.3V/0.42A, 24V to 3.3V/0.32A, 12V to 3.3V/0.22A
48V to 5V/0.38A, 24V to 5V/0.27A, 12V to 5V/0.17A
750312365 300 1.8 4:1 Wurth Elektronik 48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A
750312558 300 1.75 2:1:1 Wurth Elektronik 48V to ±12V/67mA, 24V to ±12V/50mA, 12V to ±12V/33mA
48V to ±15V/62mA, 24V to ±15V/44mA, 12V to ±15V/28mA
750312559 300 2 1:1 Wurth Elektronik 48V to 24V/67mA, 24V to 24V/50mA, 12V to 24V/33mA
750311019 400 5 6:1:2 Wurth Elektronik 48V to 3.3V/0.42A, 24V to 3.3V/0.32A, 12V to 3.3V/0.22A
48V to 5V/0.38A, 24V to 5V/0.27A, 12V to 5V/0.17A
750311558 300 1.5 4:1:1 Wurth Elektronik 48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A
750311660 350 3 2:1:0.33 Wurth Elektronik 48V to 12V/0.134A, 24V to 12V/0.1A, 12V to 12V/0.066A
48V to 15V/0.124A, 24V to 15V/0.088A, 12V to 15V/0.056A
750311838 350 3 2:1:1 Wurth Elektronik 48V to ±12V/67mA, 24V to ±12V/50mA, 12V to ±12V/33mA
48V to ±15V/62mA, 24V to ±15V/44mA, 12V to ±15V/28mA
750311659 300 2 1:1:0.2 Wurth Elektronik 48V to 24V/67mA, 24V to 24V/50mA, 12V to 24V/33mA
10396-T026 300 2.5 6:1:2 Sumida 48V to 3.3V/0.42A, 24V to 3.3V/0.32A, 12V to 3.3V/0.22A
48V to 5V/0.38A, 24V to 5V/0.27A, 12V to 5V/0.17A
10396-T024 300 2 4:1:1 Sumida 48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A
10396-T022 300 2 2:1:0.33 Sumida 48V to 12V/0.134A, 24V to 12V/0.1A, 12V to 12V/0.066A
48V to 15V/0.124A, 24V to 15V/0.088A, 12V to 15V/0.056A
10396-T028 300 2.5 2:1:1 Sumida 48V to ±12V/67mA, 24V to ±12V/50mA, 12V to ±12V/33mA
48V to ±15V/62mA, 24V to ±15V/44mA, 12V to ±15V/28mA
L10-0116 500 7.3 6:1 BH Electronics 48V to 3.3V/0.42A, 24V to 3.3V/0.32A, 12V to 3.3V/0.22A
48V to 5V/0.38A, 24V to 5V/0.27A, 12V to 5V/0.17A
L10-0112 230 3.38 4:1 BH Electronics 48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A
L11-0067 230 2.16 4:1 BH Electronics 48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A
* All the transformers are rated for 1.5kV Isolation.
LT8300 VUUT 12
LT8300
12
Rev. A
For more information www.analog.com
Typically, choose the transformer turns ratio to maximize
available output power. For low output voltages (3.3V or
5V), a larger N:1 turns ratio can be used with multiple pri-
mary windings relative to the secondary to maximize the
transformer’s current gain (and output power). However,
remember that the SW pin sees a voltage that is equal
to the maximum input supply voltage plus the output
voltage multiplied by the turns ratio. In addition, leakage
inductance will cause a voltage spike (VLEAKAGE) on top of
this reflected voltage. This total quantity needs to remain
below the 150V absolute maximum rating of the SW pin to
prevent breakdown of the internal power switch. Together
these conditions place an upper limit on the turns ratio,
NPS, for a given application. Choose a turns ratio low
enough to ensure:
NPS <
150V
V
IN(MAX)
V
LEAKAGE
V
OUT
+V
F
For lower output power levels, choose a smaller N:1 turns
ratio to alleviate the SW pin voltage stress. Although a
1:N turns ratio makes it possible to have very high output
voltages without exceeding the breakdown voltage of the
internal power switch, the multiplied parasitic capacitance
through turns ratio coupled with the relatively resistive
150V internal power switch may cause the switch turn-
on current spike ringing beyond 160ns leading-edge
blanking, thereby producing light load instability in cer-
tain applications. So any 1:N turns ratio should be fully
evaluated before its use with the LT8300.
The turns ratio is an important element in the isolated
feedback scheme, and directly affects the output voltage
accuracy. Make sure the transformer manufacturer speci-
fies turns ratio accuracy within ±1%.
APPLICATIONS INFORMATION
Saturation Current
The current in the transformer windings should not exceed
its rated saturation current. Energy injected once the core
is saturated will not be transferred to the secondary and
will instead be dissipated in the core. When designing
custom transformers to be used with the LT8300, the
saturation current should always be specified by the
transformer manufacturers.
Winding Resistance
Resistance in either the primary or secondary windings
will reduce overall power efficiency. Good output volt
-
age regulation will be maintained independent of winding
resistance due to the boundary/discontinuous conduction
mode operation of the LT8300.
Leakage Inductance and Snubbers
Transformer leakage inductance on either the primary
or secondary causes a voltage spike to appear on the
primary after the power switch turns off. This spike is
increasingly prominent at higher load currents where
more stored energy must be dissipated. It is very impor-
tant to minimize transformer leakage inductance.
When designing an application, adequate margin should
be kept for the worst-case leakage voltage spikes even
under overload conditions. In most cases shown in
Figure5, the reflected output voltage on the primary plus
VIN should be kept below 120V. This leaves at least 30V
margin for the leakage spike across line and load condi-
tions. A larger voltage margin will be required for poorly
wound transformers or for excessive leakage inductance.
In addition to the voltage spikes, the leakage inductance
also causes the SW pin ringing for a while after the power
switch turns off. To prevent the voltage ringing falsely trig-
ger boundary mode detector, the LT8300 internally blanks
the boundary mode detector for approximately 250ns.
Any remaining voltage ringing after 250ns may turn the
power switch back on again before the secondary cur-
rent falls to zero. So the leakage inductance spike ringing
should be limited to less than 250ns.
LT8300 13
LT8300
13
Rev. A
For more information www.analog.com
APPLICATIONS INFORMATION
A snubber circuit is recommended for most applications.
Two types of snubber circuits shown in Figure6 that can
protect the internal power switch include the DZ (diode-
Zener) snubber and the RC (resistor-capacitor) snubber.
The DZ snubber ensures well defined and consistent
clamping voltage and has slightly higher power efficiency,
while the RC snubber quickly damps the voltage spike
ringing and provides better load regulation and EMI per-
formance. Figure5 shows the flyback waveforms with the
DZ and RC snubbers.
For the DZ snubber, proper care must be taken when
choosing both the diode and the Zener diode. Schottky
diodes are typically the best choice, but some PN diodes
can be used if they turn on fast enough to limit the leakage
inductance spike. Choose a diode that has a reverse-volt-
age rating higher than the maximum SW pin voltage. The
Zener diode breakdown voltage should be chosen to bal-
ance power loss and switch voltage protection. The best
compromise is to choose the largest voltage breakdown.
Use the following equation to make the proper choice:
VZENER(MAX) ≤ 150V – VIN(MAX)
For an application with a maximum input voltage of 72V,
choose a 68V Zener diode, the VZENER(MAX) of which is
around 72V and below the 78V maximum.
The power loss in the clamp will determine the power
rating of the Zener diode. Power loss in the clamp is
highest at maximum load and minimum input voltage.
The switch current is highest at this point along with the
energy stored in the leakage inductance. A 0.5W Zener
will satisfy most applications when the highest VZENER
is chosen.
Figure5. Maximum Voltages for SW Pin Flyback Waveform
Figure6. Snubber Circuits
8300 F05
VSW
tOFF > 350ns
VLEAKAGE
tSP < 250ns
VSW VSW
TIME
No Snubber with DZ Snubber with RC Snubber
tOFF > 350ns
VLEAKAGE
tSP < 250ns
TIME
tOFF > 350ns
VLEAKAGE
tSP < 250ns
TIME
<150V
<120V
<150V
<120V
<150V
<120V
8300 F06b8300 F06a
DZ Snubber RC Snubber
L
Z
D
C
R
L
LT8300 SNUBBER PERIOD 2 LPAR CPAR 14 2 R2 ws. M—< '___i}i_="">
LT8300
14
Rev. A
For more information www.analog.com
APPLICATIONS INFORMATION
Tables 2 and 3 show some recommended diodes and
Zener diodes.
Table2. Recommended Zener Diodes
PART
VZENER
(V)
POWER
(W) CASE VENDOR
MMSZ5266BT1G 68 0.5 SOD-123 On Semi
MMSZ5270BT1G 91 0.5 SOD-123
CMHZ5266B 68 0.5 SOD-123 Central
Semiconductor
CMHZ5267B 75 0.5 SOD-123
BZX84J-68 68 0.5 SOD323F NXP
BZX100A 100 0.5 SOD323F
Table3. Recommended Diodes
PART I (A)
VREVERSE
(V) CASE VENDOR
BAV21W 0.625 200 SOD-123 Diodes Inc.
BAV20W 0.625 150 SOD-123
The recommended approach for designing an RC snub-
ber is to measure the period of the ringing on the SW
pin when the power switch turns off without the snub-
ber and then add capacitance (starting with 100pF) until
the period of the ringing is 1.5 to 2 times longer. The
change in period will determine the value of the parasitic
capacitance, from which the parasitic inductance can be
determined from the initial period, as well. Once the value
of the SW node capacitance and inductance is known, a
series resistor can be added to the snubber capacitance
to dissipate power and critically dampen the ringing. The
equation for deriving the optimal series resistance using
the observed periods ( tPERIOD and tPERIOD(SNUBBED)) and
snubber capacitance (CSNUBBER) is:
CPAR =CSNUBBER
tPERIOD(SNUBBED)
tPERIOD
2
1
LPAR =tPERIOD 2
CPAR • 4π2
RSNUBBER =LPAR
CPAR
Figure7. Undervoltage Lockout (UVLO)
LT8300
GND
EN/UVLO
R1
RUN/STOP
CONTROL
(OPTIONAL)
R2
V
IN
8300 F07
Note that energy absorbed by the RC snubber will be
converted to heat and will not be delivered to the load.
In high voltage or high current applications, the snubber
may need to be sized for thermal dissipation.
Undervoltage Lockout (UVLO)
A resistive divider from VIN to the EN/UVLO pin imple-
ments undervoltage lockout (UVLO). The EN/UVLO pin
falling threshold is set at 1.223V with 16mV hysteresis.
In addition, the EN/UVLO pin sinks 2.5µA when the volt-
age at the pin is below 1.223V. This current provides user
programmable hysteresis based on the value of R1. The
programmable UVLO thresholds are:
VIN(UVLO+)=1.239V •(R1+R2)
R2 +2.A •R1
VIN(UVLO)=1.223V •(R1+R2)
R2
Figure7 shows the implementation of external shutdown
control while still using the UVLO function. The NMOS
grounds the EN/UVLO pin when turned on, and puts the
LT8300 in shutdown with quiescent current less than 2µA.
LT8300 12V+0.3V 15
LT8300
15
Rev. A
For more information www.analog.com
Minimum Load Requirement
The LT8300 samples the isolated output voltage from
the primary-side flyback pulse waveform. The flyback
pulse occurs once the primary switch turns off and the
secondary winding conducts current. In order to sample
the output voltage, the LT8300 has to turn on and off at
least for a minimum amount of time and with a minimum
frequency. The LT8300 delivers a minimum amount of
energy even during light load conditions to ensure accu-
rate output voltage information. The minimum energy
delivery creates a minimum load requirement, which can
be approximately estimated as:
ILOAD(MIN) =LPRI ISW(MIN) 2• fMIN
2 • V
OUT
LPRI = Transformer primary inductance
ISW(MIN) = Minimum switch current limit = 52mA
fMIN = Minimum switching frequency = 7.5kHz
The LT8300 typically needs less than 0.5% of its full output
power as minimum load. Alternatively, a Zener diode with
its breakdown of 20% higher than the output voltage can
serve as a minimum load if pre-loading is not acceptable.
For a 5V output, use a 6V Zener with cathode connected to
the output.
Output Short Protection
When the output is heavily overloaded or shorted, the
reflected SW pin waveform rings longer than the inter-
nal blanking time. After the 350ns minimum switch-off
time, the excessive ring falsely trigger the boundary mode
detector and turn the power switch back on again before
the secondary current falls to zero. Under this condition,
the LT8300 runs into continuous conduction mode at
750kHz maximum switching frequency. Depending on the
VIN supply voltage, the switch current may run away and
exceed 260mA maximum current limit. Once the switch
current hits 520mA over current limit, a soft-start cycle
initiates and throttles back both switch current limit and
switch frequency. This output short protection prevents
the switch current from running away and limits the aver-
age output diode current.
APPLICATIONS INFORMATION
Design Example
Use the following design example as a guide to design
applications for the LT8300. The design example involves
designing a 12V output with a 120mA load current and an
input range from 36V to 72V.
VIN(MIN) = 36V, VIN(NOM) = 48V, VIN(MAX) = 72V,
VOUT = 12V, IOUT = 120mA
Step 1: Select the Transformer Turns Ratio.
NPS <150V VIN(MAX) VLEAKAGE
VOUT +VF
VLEAKAGE = Margin for transformer leakage spike = 30V
VF = Output diode forward voltage = ~0.3V
Example:
NPS <150V 72V 30V
12V +0.3V
=3.9
The choice of transformer turns ratio is critical in deter-
mining output current capability of the converter. Table4
shows the switch voltage stress and output current capa-
bility at different transformer turns ratio.
Table4. Switch Voltage Stress and Output Current Capability
vs Turns Ratio
NPS
VSW(MAX) at
VIN(MAX) (V)
IOUT(MAX) at
VIN(MIN) (mA) DUTY CYCLE (%)
1:1 84.3 84 15-25
2:1 96.6 135 25-41
3:1 108.9 168 34-51
Since both NPS = 2 and NPS = 3 can meet the 120mA
output current requirement, NPS = 2 is chosen in this
example to allow more margin for transformer leakage
inductance voltage spike.
LT8300 LPR‘ °'sw LPm "sw 16
LT8300
16
Rev. A
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APPLICATIONS INFORMATION
Step 2: Determine the Primary Inductance.
Primary inductance for the transformer must be set above
a minimum value to satisfy the minimum switch-off and
switch-on time requirements:
LPRI tOFF(MIN) NPS • VOUT +VF
( )
ISW(MIN)
LPRI tON(MIN) • VIN(MAX)
ISW(MIN)
tOFF(MIN) = 350ns
tON(MIN) = 160ns
ISW(MIN) = 52mA
Example:
LPRI
350ns • 2 •(12V
+
0.3V)
52mA =166µH
LPRI 160ns • 72V
52mA =222µH
Most transformers specify primary inductance with a tol-
erance of ±20%. With other component tolerance consid-
ered, choose a transformer with its primary inductance
20% to 40% larger than the minimum values calculated
above. LPRI = 300µH is then chosen in this example.
Once the primary inductance has been determined, the
maximum load switching frequency can be calculated as:
fSW =1
tON +tOFF
=1
LPRI ISW
VIN
+LPRI ISW
NPS (VOUT +VF)
ISW =VOUT IOUT • 2
η• VIN D
Example:
D=(12V +0.3V) • 2
(12V +0.3V) • 2+48V =0.34
ISW =12V • 0.12A • 2
0.85 48V • 0.34 =0.21A
fSW =260kHz
The transformer also needs to be rated for the correct
saturation current level across line and load conditions. A
saturation current rating larger than 400mA is necessary
to work with the LT8300. The 10396-T022 from Sumida
is chosen as the flyback transformer.
Step 3: Choose the Output Diode.
Two main criteria for choosing the output diode include
forward current rating and reverse voltage rating. The
maximum load requirement is a good first-order guess
as the average current requirement for the output diode.
A conservative metric is the maximum switch current limit
multiplied by the turns ratio,
IDIODE(MAX) = ISW(MAX) • NPS
Example:
IDIODE(MAX) = 0.52A
Next calculate reverse voltage requirement using maxi-
mum VIN:
VREVERSE =VOUT +VIN(MAX)
N
PS
Example:
VREVERSE =12V +72V
2
=48V
The SBR0560S1 (0.5A, 60V diode) from Diodes Inc.
ischosen.
LT8300 2-12V-0.12V 17
LT8300
17
Rev. A
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Step 4: Choose the Output Capacitor.
The output capacitor should be chosen to minimize the
output voltage ripple while considering the increase in size
and cost of a larger capacitor. Use the equation below to
calculate the output capacitance:
COUT =LPRI ISW 2
2 • V
OUT
ΔV
OUT
Example:
Design for output voltage ripple less than 1% of VOUT,
i.e., 120mV.
COUT =30H •(0.21A)2
2 12V 0.12V
=4.F
Remember ceramic capacitors lose capacitance with
applied voltage. The capacitance can drop to 40% of
quoted capacitance at the maximum voltage rating. So a
10uF, 16V rating ceramic capacitor is chosen.
Step 5: Design Snubber Circuit.
The snubber circuit protects the power switch from leak-
age inductance voltage spike. A DZ snubber is recom-
mended for this application because of lower leakage
inductance and larger voltage margin. The Zener and the
diode need to be selected.
The maximum Zener breakdown voltage is set according
to the maximum VIN:
VZENER(MAX) ≤ 150V – VIN(MAX)
Example:
VZENER(MAX) ≤ 150V – 72V = 78V
APPLICATIONS INFORMATION
A 68V Zener with a maximum of 72V will provide optimal
protection and minimize power loss. So a 68V, 0.5W Zener
from On Semiconductor (MMSZ5266BT1G) is chosen.
Choose a diode that is fast and has sufficient reverse volt-
age breakdown:
VREVERSE > VSW(MAX)
VSW(MAX) = VIN(MAX) + VZENER(MAX)
Example:
VREVERSE > 144V
A 150V, 0.6A diode from Diodes Inc. (BAV20W) is chosen.
Step 6: Select the RFB Resistor.
Use the following equation to calculate the starting value
for RFB:
RFB =NPS (VOUT +VF)
100µA
Example:
RFB =2 (12V +0.3V)
100µA =246k
Depending on the tolerance of standard resistor values,
the precise resistor value may not exist. For 1% standard
values, a 243k resistor in series with a 3.01k resistor
should be close enough. As discussed in the Application
Information section, the final R
FB
value should be adjusted
on the measured output voltage.
LT8300 2-12V R2 18
LT8300
18
Rev. A
For more information www.analog.com
APPLICATIONS INFORMATION
Step 7: Select the EN/UVLO Resistors.
Determine the amount of hysteresis required and calcu-
late R1 resistor value:
VIN(HYS) = 2.5µA • R1
Example:
Choose 2.5V of hysteresis,
R1 = 1M
Determine the UVLO thresholds and calculate R2
resistorvalue:
VIN(UVLO+)=1.239V •(R1+R2)
R2
+2.5µA •R1
Example:
Set VIN UVLO rising threshold to 34.5V,
R2 = 40.2k
VIN(UVLO+) = 34.1V
VIN(UVLO–) = 31.6V
Step 8: Ensure minimum load.
The theoretical minimum load can be approximately esti-
mated as:
ILOAD(MIN) =30H •(52mA)2• 7.5kHz
2 12V
=0.25mA
Remember to check the minimum load requirement in real
application. The minimum load occurs at the point where
the output voltage begins to climb up as the converter
delivers more energy than what is consumed at the out-
put. The real minimum load for this application is about
0.6mA, 0.5% of 120mA maximum load. In this example,
a 20k resistor is selected as the minimum load.
LT8300 19
LT8300
19
Rev. A
For more information www.analog.com
TYPICAL APPLICATIONS
LT8300
T1
6:1
D1
RFB
SW
300µH 8µH
EN/UVLO
1M
2.2µF
40.2k
VIN
VIN
36V TO 72V
VOUT
+
5V
1mA TO 330mA
VOUT
GND
316k
T1: WURTH 750312557
D1: DIODES INC. SBR2A30P1
47µF
8300 TA02
5V Micropower Isolated Flyback Converter
12V Micropower Isolated Flyback Converter
LT8300
T1
2:1
D1
RFB
SW
300µH 75µH
EN/UVLO
1M
2.2µF
40.2k
VIN
VIN
36V TO 72V
VOUT+
12V
0.6mA TO 120mA
VOUT
GND
243k
T1: SUMIDA 10396-TO22
D1: DIODES INC. SBR0560S1
10µF
8300 TA03
LT8300 +1 I~ +1 I~
LT8300
20
Rev. A
For more information www.analog.com
TYPICAL APPLICATIONS
LT8300
T1
8:1
D1
RFB
SW
400µH 6µH
EN/UVLO
1M
2.2µF
40.2k
VIN
VIN
36V TO 72V
VOUT+
3.3V
2mA TO 440mA
VOUT
GND
287k
T1: WURTH 750312367
D1: NXP PMEG2020EH
100µF
8300 TA05
24V Micropower Isolated Flyback Converter
3.3V Micropower Isolated Flyback Converter
LT8300
T1
1:1
D1
RFB
SW
300µH 300µH
EN/UVLO
1M
2.2µF
40.2k
VIN
VIN
36V TO 72V
VOUT+
24V
0.3mA TO 60mA
VOUT
GND
243k
T1: WURTH 750311559
D1: DIODES DFLS 1200-7
4.7µF
8300 TA04
LT8300 21
LT8300
21
Rev. A
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TYPICAL APPLICATIONS
VIN to (VIN + 10V) Micropower Converter
VIN to (VIN – 10V) Micropower Converter
LT8300 D1
Z1
RFB
SW
L1
330µH
EN/UVLO
1M
1µF
118k
VIN
VIN
15V TO 80V VOUT+
10V
100mA
VOUT
GND
102k
L1: COILTRONICS DR73-331-R
D1: DIODES INC. SBR1U150SA
Z1: CENTRAL CMDZ12L
8300 TA07
4.7µF
LT8300 D1
Z1
RFB
SW
L1
330µH
EN/UVLO
1M
1µF
118k
VIN
VIN
15V TO 80V
VOUT+
10V
50mA
VOUT
GND
102k
L1: COILTRONICS DR73-331-R
D1: DIODES INC. SBR1U150SA
Z1: CENTRAL CMDZ12L
8300 TA06
4.7µF
LT8300 Em; r L LEW o L EL L NOTE V DLMENSLCNS ARE LN MLLL 2 DRAWWG NOT TO SCALE 3 DLMENSLCNS ARE LNCLUSIVE OF PLATING 4 DLMENSLCNS ARE ExCLuSLonEMULDEL 5 MOLD FLASH SHALL NOT EXCEED 0 254m 6 JEDEC PACKAGE REFERENCE Ls M0493 #L Le 4»
LT8300
22
Rev. A
For more information www.analog.com
PACKAGE DESCRIPTION
S5 Package
5-Lead Plastic TSOT-23
(Reference LTC DWG # 05-08-1635 Rev B)
1.50 – 1.75
(NOTE 4)
2.80 BSC
0.30 – 0.45 TYP
5 PLCS (NOTE 3)
DATUM ‘A’
0.09 – 0.20
(NOTE 3) S5 TSOT-23 0302 REV B
PIN ONE
2.90 BSC
(NOTE 4)
0.95 BSC
1.90 BSC
0.80 0.90
1.00 MAX 0.01 – 0.10
0.20 BSC
0.30 – 0.50 REF
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. JEDEC PACKAGE REFERENCE IS MO-193
3.85 MAX
0.62
MAX
0.95
REF
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
1.4 MIN
2.62 REF
1.22 REF
LT8300 23
LT8300
23
Rev. A
For more information www.analog.com
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog
Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications
subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
REVISION HISTORY
REV DATE DESCRIPTION PAGE NUMBER
A 06/19 Added Automotive Part Numbers 2
LT8300 ANALOG DEVICES
LT8300
24
Rev. A
For more information www.analog.com
ANALOG DEVICES, INC. 2012-2019
06/19
www.analog.com
RELATED PARTS
TYPICAL APPLICATION
3.3V Isolated Converter (Conforming to DEF-STAN61-5)
PART NUMBER DESCRIPTION COMMENTS
LT3511/LT3512 100V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 240mA/420mA Switch,
MSOP-16(12)
LT3748 100V Isolated Flyback Controller 5V ≤ VIN ≤ 100V, No Opto Flyback , MSOP-16 with High Voltage Spacing
LT3798 Off-Line Isolated No Opto-Coupler Flyback Controller
with Active PFC
VIN and VOUT Limited Only by External Components
LT3573/LT3574/LT3575 40V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 1.25A/0.65A/2.5A Switch
LT3757/LT3759/LT3758 40V/100V Flyback/Boost Controllers Universal Controllers with Small Package and Powerful Gate Drive
LT3957/LT3958 40V/100V Flyback/Boost Converters Monolithic with Integrated 5A/3.3A Switch
LTC3803/LTC3803-3/
LTC3803-5
200kHz/300kHz Flyback Controllers in SOT-23 VIN and VOUT Limited by External Components
LTC3805/LTC3805-5 Adjustable Frequency Flyback Controllers VIN and VOUT Limited by External Components
VIN (V)
IVIN (µA)
400
200
300
100
0
8300 TA08b
323028262420 2218
LT8300
L1
1:1
D1
RFB
SW
150µH 150µH
EN/UVLO
1M
1µF
93.1k
VIN
VIN
18V TO 32V
VOUT+
3.3V
0mA TO 20mA
VOUT
GND
42.2k D1: DIODES INC. SBR0560S1-7
L1: DRQ73-151-R
Z1: CENTRAL CMDZ4L7
1µF 1µF
Z1
8300 TA08a
LT3009-3.3
GND
SHDN
IN OUT
Input Current with No Load

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