LT3475/-1 Datasheet by Analog Devices Inc.

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L7Llnemww TECHNOLOGY [J >1 I TL T ‘i’ J_—— J'WA JMTE 49TH; L7 LJUW
LT3475/LT3475-1
1
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Dual Step-Down
1.5A LED Driver
The LT®3475/LT3475-1 are dual step-down DC/DC
converters designed to operate as a constant-current
source. An internal sense resistor monitors the output
current allowing accurate current regulation ideal for
driving high current LEDs. The high side current sense al-
lows grounded cathode LED operation. High output current
accuracy is maintained over a wide current range, from
50mA to 1.5A, allowing a wide dimming range. Unique
PWM circuitry allows a dimming range of 3000:1, avoid-
ing the color shift normally associated with LED current
dimming.
The high switching frequency offers several advantages,
permitting the use of small inductors and ceramic capaci-
tors. Small inductors combined with the 20 lead TSSOP
surface mount package save space and cost versus
alternative solutions. The constant switching frequency
combined with low-impedance ceramic capacitors result
in low, predictable output ripple.
With its wide input range of 4V to 36V, the LT3475/LT3475-1
regulate a broad array of power sources. A current mode
PWM architecture provides fast transient response and
cycle-by-cycle current limiting. Frequency foldback and
thermal shutdown provide additional protection.
Automotive and Avionic Lighting
Architectural Detail Lighting
Display Backlighting
Constant-Current Sources
True Color PWMTM Delivers Constant Color with
3000:1 Dimming Range
Wide Input Range: 4V to 36V Operating, 40V
Maximum
Accurate and Adjustable Control of LED Current
from 50mA to 1.5A
High Side Current Sense Allows Grounded Cathode
LED Operation
Open LED (LT3475) and Short Circuit Protection
LT3475-1 Drives LED Strings Up to 25V
Accurate and Adjustable 200kHz to 2MHz
Switching Frequency
Anti-Phase Switching Reduces Ripple
Uses Small Inductors and Ceramic Capacitors
Available in the Compact 20-Lead TSSOP Thermally
Enhanced Surface Mount Package
Dual Step-Down 1.5A LED Driver
APPLICATIO S
U
FEATURES DESCRIPTIO
U
TYPICAL APPLICATIO
U
Effi ciency
, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners. Patents Pending.
BOOST1 BOOST2
SW1 SW2
OUT1 OUT2
LED1 LED2
PWM1 PWM2
VC1 VC2
REF RT
VADJ1 VADJ2
GND
LT3475
VIN
VIN
5V TO 36V
SHDN
0.22μF
4.7μF
10μH 10μH
0.22μF
24.3k
2.2μF 2.2μF
DIMMING*
CONTROL
*DIMMING
CONTROL
0.1μF
0.1μF
3475 TA01
1.5A LED
CURRENT
1.5A LED
CURRENT
fSW = 600kHz*SEE APPLICATIONS SECTION FOR DETAILS
LED CURRENT (A)
0
EFFICIENCY (%)
70
75
80
3475 TA01b
65
60
55 0.5 1
85
90
95
1.5
TWO SERIES CONNECTED
WHITE 1.5A LEDS
SINGLE WHITE 1.5A LED
VIN = 12V
LT3475/LT3475—1 3333333333 EEEEEEEEKE
LT3475/LT3475-1
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VIN Pin .........................................................(-0.3V), 40V
BOOST Pin Voltage ...................................................60V
BOOST Above SW Pin ...............................................30V
OUT, LED, Pins (LT3475) ...........................................15V
OUT, LED Pins (LT3475-1) .........................................25V
PWM Pin ...................................................................15V
VADJ Pin ......................................................................6V
VC, RT
, REF Pins ..........................................................3V
SHDN Pin ...................................................................VIN
Maximum Junction Temperature (Note 2)............. 125°C
Operating Temperature Range (Note 3)
LT3475E/LT3475E-1 ............................. –40°C to 85°C
LT3475I/LT3475I-1 ............................. –40°C to 125°C
Storage Temperature Range ................... –65°C to 150°C
Lead Temperature Range (Soldering, 10 sec) ....... 300°C
(Note 1)
The
denotes the specifi cations which apply over the full operating
temperature range, otherwise specifi cations are at TA = 25°C. VIN = 12V, VBOOST = 16V, VOUT = 4V unless otherwise noted (Note 3)
PARAMETER CONDITIONS MIN TYP MAX UNITS
Minimum Input Voltage 3.7 4 V
Input Quiescent Current Not Switching 6 8 mA
Shutdown Current
SHDN = 0.3V, VBOOST = VOUT = 0V
0.01 2 μA
ELECTRICAL CHARACTERISTICS
ABSOLUTE AXI U RATI GS
W
WW
U
FE PACKAGE
20-LEAD PLASTIC TSSOP
1
2
3
4
5
6
7
8
9
10
TOP VIEW
20
19
18
17
16
15
14
13
12
11
OUT1
LED1
BOOST1
SW1
VIN
VIN
SW2
BOOST2
LED2
OUT2
PWM1
VADJ1
VC1
REF
SHDN
GND
RT
VC2
VADJ2
PWM2
21
TJMAX = 125°C, θJA = 30°C/W, θJC = 8°C/W
EXPOSED PAD (PIN 21) IS GROUND AND MUST
BE ELECTRICALLY CONNECTED TO THE PCB.
LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE
LT3475EFE#PBF LT3475EFE#TRPBF LT3475EFE 20-Lead Plastic TSSOP –40°C to 85°C
LT3475IFE#PBF LT3475IFE#TRPBF LT3475IFE 20-Lead Plastic TSSOP –40°C to 125°C
LT3475EFE-1#PBF LT3475EFE-1#TRPBF LT3475FE-1 20-Lead Plastic TSSOP –40°C to 85°C
LT3475IFE-1#PBF LT3475IFE-1#TRPBF LT3475FE-1 20-Lead Plastic TSSOP –40°C to 125°C
Consult LTC Marketing for parts specifi ed with wider operating temperature ranges. *The temperature grade is identifi ed by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based fi nish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifi cations, go to: http://www.linear.com/tapeandreel/
PIN CONFIGURATION
ORDER INFORMATION
LT3475/ LT3475—1 L7 LJUW
LT3475/LT3475-1
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Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: This IC includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed 125°C when overtemperature protection is active.
Continuous operation above the specifi ed maximum operating junction
temperature may impair device reliability.
Note 3: The LT3475E and LT3475E-1 are guaranteed to meet performance
specifi cations from 0°C to 85°C. Specifi cations over the –40°C to 85°C
operating temperature range are assured by design, characterization and
correlation with statistical process controls. The LT3475I and LT3475I-1
are guaranteed to meet performance specifi cations over the –40°C to
125°C operating temperature range.
Note 4: Current fl ows out of pin.
Note 5: Current fl ows into pin.
Note 6: Current limit is guaranteed by design and/or correlation to static
test. Slope compensation reduces current limit at higher duty cycles.
The denotes the specifi cations which apply over the full operating
temperature range, otherwise specifi cations are at TA = 25°C. VIN = 12V, VBOOST = 16V, VOUT = 4V unless otherwise noted (Note 3)
PARAMETER CONDITIONS MIN TYP MAX UNITS
LED Pin Current VADJ Tied to VREF • 2/3
VADJ Tied to VREF • 7/30
LT3475E/LT3475E-1 0°C to 85°C
0.97
0.94
0.336
0.325
0.31
1.00
0.350
1.03
1.04
0.364
0.375
0.385
A
A
A
A
A
REF Voltage 1.22 1.25 1.27 V
Reference Voltage Line Regulation 4V < VIN < 40V 0.05 %/V
Reference Voltage Load Regulation 0 < IREF < 500μA 0.0002 %/μA
VADJ Pin Bias Current (Note 4) 40 400 nA
Switching Frequency RT = 24.3k 530 600 640 kHz
Maximum Duty Cycle RT = 24.3k
RT = 4.32k
RT = 100k
90 95
80
98
%
%
%
Switching Phase RT = 24.3k 150 180 210 Deg
Foldback Frequency RT = 24.3k, VOUT = 0V 80 kHz
SHDN
Threshold (to Switch) 2.5 2.6 2.74 V
SHDN
Pin Current (Note 5) VSHDN =
2.6V
7911 μA
PWM Threshold 0.3 0.8 1.2 V
VC Switching Threshold 0.8 V
VC Source Current VC = 1V 50 μA
VC Sink Current VC = 1V 50 μA
LED to VC Transresistance 500 V/A
LED to VC Current Gain 1 mA/μA
VC to Switch Current Gain 2.6 A/V
VC Clamp Voltage 1.8 V
VC Pin Current in PWM Mode VC = 1V, VPWM = 0.3V 10 400 nA
OUT Pin Clamp Voltage (LT3475) 13.5 14 14.5 V
OUT Pin Current in PWM Mode VOUT = 4V, VPWM = 0.3V 25 50 μA
Switch Current Limit (Note 6) 2.3 2.7 3.2 A
Switch VCESAT ISW =1.5A 350 500 mV
BOOST Pin Current ISW =1.5A 25 40 mA
Switch Leakage Current 0.1 10 μA
Minimum Boost Voltage Above SW 1.8 2.5 V
ELECTRICAL CHARACTERISTICS
LT3475/LT3475—1 L7LJCUEN2
LT3475/LT3475-1
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Switch Current Limit
vs Duty Cycle
Switch Current Limit vs
Temperature
Oscillator Frequency
vs Temperature Oscillator Frequency Foldback Oscillator Frequency vs RT
Current Limit vs Output Voltage
TYPICAL PERFOR A CE CHARACTERISTICS
UW
LED Current vs VADJ LED Current vs Temperature Switch On Voltage
VADJ (V)
0
0
LED CURRENT (A)
0.25
0.50
0.75
1.00
1.25
1.50
0.25 0.5 0.75 1
3475 G01
1.25
TA = 25°C
TEMPERATURE (˚C)
–50
LED CURRENT (A)
0.8
1.0
1.2
25 75
3475 G02
0.6
0.4
–25 0 50 100 125
0.2
0
VADJ = VREF • 7/30
VADJ = VREF • 2/3
SWITCH CURRENT (A)
0
0
SWITCH ON VOLTAGE (mV)
100
200
300
400
0.5 1.0 1.5 2.0
3475 G03
500
600 TA = 25°C
DUTY CYCLE (%)
0
0
CURRENT LIMIT (A)
0.5
1.0
1.5
2.0
2.5
3.0
20 40 60 80
3475 G04
100
MINIMUM
TYPICAL
TA = 25°C
TEMPERATURE (°C)
–50
2.0
2.5
3.5
25 75
3475 G05
1.5
1.0
–25 0 50 100 125
0.5
0
3.0
CURRENT LIMIT (A)
VOUT (V)
0
0
CURRENT LIMIT (A)
0.5
1.0
1.5
2.0
0.5 2.5 3.0 3.52.01.0 1.5 4.0
3475 G06
2.5
3.0 TA = 25°C
TEMPERATURE (˚C)
–50
OSCILLATOR FREQUENCY (kHz)
600
650
700
25 75
3475 G07
550
500
–25 0 50 100 125
450
400
RT = 24.3kΩ
VOUT (V)
0
500
600
700
2.0
3475 G08
400
300
0.5 1.0 1.5 2.5
200
100
0
OSCILLATOR FREQUENCY (kHz)
TA = 25°C
RT = 24.3kΩ
RT (kΩ)
1
10
OSCILLATOR FREQUENCY (kHz)
1000
10 100
3475 G09
TA = 25°C
LT3475/ LT3475—1 LT3475 mum / OUTPUT L7 LJUW
LT3475/LT3475-1
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PI FU CTIO S
UUU
Boost Pin Current Quiescent Current
OUT1, OUT2 (Pins 1, 10): The OUT pin is the input to the
current sense resistor. Connect this pin to the inductor
and the output capacitor.
LED1, LED2 (Pins 2, 9): The LED pin is the output of
the current sense resistor. Connect the anode of the LED
here.
VIN (Pins 5, 6): The VIN pins supply current to the internal
circuitry and to the internal power switches and must be
locally bypassed.
SW1, SW2 (Pins 4, 7): The SW pin is the output of the
internal power switch. Connect this pin to the inductor,
switching diode and boost capacitor.
BOOST1, BOOST2 (Pins 3, 8): The BOOST pin is used to
provide a drive voltage, higher than the input voltage, to
the internal bipolar NPN power switch.
GND (Pins 15, Exposed Pad Pin 21): Ground. Tie the GND
pin and the exposed pad directly to the ground plane. The
exposed pad metal of the package provides both electrical
contact to ground and good thermal contact to the printed
circuit board. The exposed pad must be soldered to the
circuit board for proper operation. Use a large ground plane
and thermal vias to optimize thermal performance.
Open-Circuit Output Voltage and
Input Current
Reference Voltage
Minimum Input Voltage, Single
1.5A White LED
Minimum Input Voltage, Two Series
Connected 1.5A White LEDs
TYPICAL PERFOR A CE CHARACTERISTICS
UW
SWITCH CURRENT (A)
0
BOOST PIN CURRENT (mA)
20
25
30
1.5
3475 G10
15
10
0.5 1.0 2.0
5
0
35 TA = 25°C
VIN (V)
0
OUTPUT VOLTAGE (V)
INPUT CURRENT (mA)
25
40
35
40
3475 G12
10
010 20 30
50
20
15
30
5
8
12
4
0
14
6
10
2
45
TA = 25°C
LT3475-1
LT3475-1
LT3475
LT3475
OUTPUT VOLTAGE
INPUT CURRENT
VIN (V)
0
INPUT CURRENT (mA)
4
5
6
30
3475 G11
3
2
10 20 40
1
0
7TA = 25°C
LED CURRENT (A)
0
0
VIN (V)
1
2
3
4
6
0.5 1
3475 G14
1.5
5
TA = 25°C
TO RUN LED VOLTAGE
TO START
TEMPERATURE (˚C)
–50
VREF (V)
1.26
1.27
1.28
25 75
3475 G13
1.25
1.24
–25 0 50 100 125
1.23
1.22
LED CURRENT (A)
0
5
VIN (V)
6
7
8
9
10
0.5 1
3475 G15
1.5
TA = 25°C
TO RUN
LED VOLTAGE
TO START
LT3475/LT3475—1 L7LJCUEN2
LT3475/LT3475-1
6
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3475 BD
QR
QS
INT REG
AND
UVLO
SLAVE
OSC
SLOPE COMP SLOPE COMP
MOSC 1 MOSC 2
FREQUENCY
FOLDBACK
FREQUENCY
FOLDBACK
+
QR
QS
MASTER
OSC
SLAVE
OSC
+
DRIVER
gm1 gm2
BOOST1 BOOST2
D1 D2
D3 D4
DLED1 DLED 2
Q1 Q2
Q3 Q4
C1 C2
C
IN
C
OUT1
C
OUT2
V
C2
C
C1
V
C1
V
ADJ1
V
ADJ2
V
IN
V
IN
V
IN
R
T
SHDN
C
C2
R
T
GND
EXPOSED
PAD
REF
PWM 1
PWM2
LED1 LED2
SW1 SW2
OUT1 OUT2
0.067Ω 100Ω 0.067Ω100Ω
1.25k
1.25V
1.25k
2V 2V
L1 L2
DRIVER
C1 C2
BLOCK DIAGRAM
RT (Pin 14): The RT pin is used to set the internal
oscillator frequency. Tie a 24.3k resistor from RT to GND
for a 600kHz switching frequency.
SHDN
(Pin 16): The
SHDN
pin is used to shut down the
switching regulator and the internal bias circuits. The
2.6V switching threshold can function as an accurate
undervoltage lockout. Pull below 0.3V to shut down the
LT3475/LT3475-1. Pull above 2.6V to enable the LT3475/
LT3475-1. Tie to VIN if the
SHDN
function is unused.
REF (Pin 17): The REF pin is the buffered output of the
internal reference. Either tie the REF pin to the VADJ pin
for a 1.5A output current, or use a resistor divider to
generate a lower voltage at the VADJ pin. Leave this pin
unconnected if unused.
VC1, VC2 (Pins 18, 13): The VC pin is the output of the
internal error amp. The voltage on this pin controls the
peak switch current. Use this pin to compensate the
control loop.
VADJ1, VADJ2 (Pins 19, 12): The VADJ pin is the input to
the internal voltage-to-current amplifi er. Connect the VADJ
pin to the REF pin for a 1.5A output current. For lower
output currents, program the VADJ pin using the following
formula: ILED = 1.5A • VADJ/1.25V.
PWM1, PWM2 (Pins 20, 11): The PWM pin controls the
connection of the VC pin to the internal circuitry. When
the PWM pin is low, the VC pin is disconnected from the
internal circuitry and draws minimal current. If the PWM
feature is unused, leave this pin unconnected.
PI FU CTIO S
UUU
LT3475/ LT3475—1 L7 LJUW
LT3475/LT3475-1
7
3475fb
The LT3475 is a dual constant frequency, current mode
regulator with internal power switches capable of gen-
erating constant 1.5A outputs. Operation can be best
understood by referring to the Block Diagram.
If the SHDN pin is tied to ground, the LT3475 is shut
down and draws minimal current from the input source
tied to VIN. If the SHDN pin exceeds 1V, the internal bias
circuits turn on, including the internal regulator, reference
and oscillator. The switching regulators will only begin to
operate when the SHDN pin exceeds 2.6V.
The switcher is a current mode regulator. Instead of directly
modulating the duty cycle of the power switch, the feedback
loop controls the peak current in the switch during each
cycle. Compared to voltage mode control, current mode
control improves loop dynamics and provides cycle-by-
cycle current limit.
A pulse from the oscillator sets the RS fl ip-fl op and turns
on the internal NPN bipolar power switch. Current in the
switch and the external inductor begins to increase. When
this current exceeds a level determined by the voltage at
VC, current comparator C1 resets the fl ip-fl op, turning
off the switch. The current in the inductor fl ows through
the external Schottky diode and begins to decrease. The
cycle begins again at the next pulse from the oscillator.
In this way, the voltage on the VC pin controls the current
through the inductor to the output. The internal error
amplifi er regulates the output current by continually
adjusting the VC pin voltage. The threshold for switching
on the VC pin is 0.8V, and an active clamp of 1.8V limits
the output current.
The voltage on the VADJ pin sets the current through the
LED pin. The NPN, Q3, pulls a current proportional to the
voltage on the VADJ pin through the 100Ω resistor. The gm
amplifi er servos the VC pin to set the current through the
0.067Ω resistor and the LED pin. When the voltage drop
across the 0.067Ω resistor is equal to the voltage drop
across the 100Ω resistor, the servo loop is balanced.
Tying the REF pin to the VADJ pin sets the LED pin current
to 1.5A. Tying a resistor divider to the REF pin allows the
programming of LED pin currents of less than 1.5A. LED
pin current can also be programmed by tying the VADJ pin
directly to a voltage source.
An LED can be dimmed with pulse width modulation
using the PWM pin and an external NFET. If the PWM
pin is unconnected or is pulled high, the part operates
nominally. If the PWM pin is pulled low, the VC pin is dis-
connected from the internal circuitry and draws minimal
current from the compensation capacitor. Circuitry draw-
ing current from the OUT pin is also disabled. This way,
the VC pin and the output capacitor store the state of
the LED pin current until the PWM is pulled high again.
This leads to a highly linear relationship between pulse
width and output light, allowing for a large and accurate
dimming range.
The RT pin allows programming of the switching frequency.
For applications requiring the smallest external components
possible, a fast switching frequency can be used. If low
dropout or very high input voltages are required, a slower
switching frequency can be programmed.
During startup VOUT will be at a low voltage. The NPN,
Q3, can only operate correctly with suffi cient voltage
of ≈1.7V at VOUT
, A comparator senses VOUT and forces
the VC pin high until VOUT rises above 2V, and Q3 is op-
erating correctly.
The switching regulator performs frequency foldback
during overload conditions. An amplifi er senses when
VOUT is less than 2V and begins decreasing the oscillator
frequency down from full frequency to 15% of the nominal
frequency when VOUT = 0V. The OUT pin is less than 2V
during startup, short circuit, and overload conditions.
Frequency foldback helps limit switch current under these
conditions.
The switch driver operates either from VIN or from the
BOOST pin. An external capacitor and Schottky diode
are used to generate a voltage at the BOOST pin that
is higher than the input supply. This allows the driver
to saturate the internal bipolar NPN power switch for
effi cient operation.
OPERATION
LT3475/LT3475—1 L7LJCUEN2
LT3475/LT3475-1
8
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Open Circuit Protection
The LT3475 has internal open-circuit protection. If the LED
is absent or is open circuit, the LT3475 clamps the voltage
on the LED pin at 14V. The switching regulator then oper-
ates at a very low frequency to limit the input current. The
LT3475-1 has no internal open circuit protection. With the
LT3475-1, be careful not to violate the ABSMAX voltage of
th BOOST pin; if VIN > 25V, external open circuit protection
circuitry (as shown in Figure 1) may be necessary.The
output voltage during an open LED condition is shown in
the Typical Performance Characteristics section.
Undervoltage Lockout
Undervoltage lockout (UVLO) is typically used in situations
where the input supply is current limited, or has high source
resistance. A switching regulator draws constant power
from the source, so the source current increases as the
source voltage drops. This looks like a negative resistance
load to the source and can cause the source to current limit
or latch low under low source voltage conditions. UVLO
prevents the regulator from operating at source voltages
where these problems might occur.
An internal comparator will force the part into shut-
down when VIN falls below 3.7V. If an adjustable UVLO
threshold is required, the SHDN pin can be used. The
threshold voltage of the SHDN pin comparator is 2.6V. An
internal resistor pulls 9μA to ground from the SHDN pin
at the UVLO threshold.
Choose resistors according to the following formula:
R2 =2.6V
VTH –2.6V
R1 –9μA
VTH = UVLO Threshold
Example: Switching should not start until the input is
above 8V.
VTH = 8V
R1=100k
R2 =2.6V
8V – 2.6V
100k –9μA
=57.6k
Figure 2. Undervoltage Lockout
GND
9μA
2.6V
VIN
VC
LT3475
R1
C1 R2
VIN
SHDN
3475 F02
Table 1. Switching Frequencies
SWITCHING FREQUENCY (MHz) RT (kΩ)
2 4.32
1.5 6.81
1.2 9.09
1 11.8
0.8 16.9
0.6 24.3
0.4 40.2
0.3 57.6
0.2 100
Keep the connections from the resistors to the SHDN pin
short and make sure the coupling to the SW and BOOST
pins is minimized. If high resistance values are used, the
SHDN pin should be bypassed with a 1nF capacitor to
prevent coupling problems from switching nodes.
Setting the Switching Frequency
The LT3475 uses a constant frequency architecture that
can be programmed over a 200kHz to 2MHz range with a
single external timing resistor from the RT pin to ground.
A graph for selecting the value of RT for a given operating
frequency is shown in the Typical Applications section.
VC
100k
10k
22V
OUT
3475 F01
Figure 1. External Overvoltage Protection
Circuitry for the LT3475-1
APPLICATIONS INFORMATION
LT3475/ LT3475—1 VOUT VF VOUT VF MMMJJ'JLHM L7 LJDW
LT3475/LT3475-1
9
3475fb
Table 1 shows suggested RT selections for a variety of
switching frequencies.
Operating Frequency Selection
The choice of operating frequency is determined by
several factors. There is a tradeoff between effi ciency and
component size. A higher switching frequency allows the
use of smaller inductors at the cost of increased switching
losses and decreased effi ciency.
Another consideration is the maximum duty cycle. In certain
applications, the converter needs to operate at a high duty
cycle in order to work at the lowest input voltage possible.
The LT3475 has a fi xed oscillator off time and a variable
on time. As a result, the maximum duty cycle increases
as the switching frequency is decreased.
Input Voltage Range
The minimum operating voltage is determined either by the
LT3475’s undervoltage lockout of 4V, or by its maximum
duty cycle. The duty cycle is the fraction of time that the
internal switch is on and is determined by the input and
output voltages:
DC =VOUT +VF
()
VIN –V
SW +VF
()
where VF is the forward voltage drop of the catch diode
(~0.4V) and VSW is the voltage drop of the internal switch
(~0.4V at maximum load). This leads to a minimum input
voltage of:
V
IN MIN
()
=VOUT +V
F
DCMAX
–V
F+VSW
with DCMAX = 1–tOFF(MIN) • f
where t0FF(MIN) is equal to 167ns and f is the switching
frequency.
Example: f = 600kHz, VOUT = 4V
DCMAX =1167ns • 600kHz =0.90
V
IN MIN
()
=4V +0.4V
0.9 – 0.4V +0.4V =4.9V
The maximum operating voltage is determined by the
absolute maximum ratings of the VIN and BOOST pins,
and by the minimum duty cycle.
V
IN MAX
()
=VOUT +V
F
DCMIN
–V
F+VSW
with DCMIN = tON(MIN) • f
where tON(MIN) is equal to 140ns and f is the switching
frequency.
Example: f = 750kHz, VOUT = 3.4V
DCMIN =140ns • 750kHz =0.105
V
IN MAX
()
=3.4V +0.4V
0.105 – 0.4V +0.4V =36V
The minimum duty cycle depends on the switching fre-
quency. Running at a lower switching frequency might
allow a higher maximum operating voltage. Note that
this is a restriction on the operating input voltage; the
circuit will tolerate transient inputs up to the Absolute
Maximum Ratings of the VIN and BOOST pins. The input
voltage should be limited to the VIN operating range (36V)
during overload conditions (short circuit or start up).
Minimum On Time
The LT3475 will regulate the output current at input volt-
ages greater than VIN(MAX). For example, an application
with an output voltage of 3V and switching frequency of
1.2MHz has a VIN(MAX) of 20V, as shown in Figure 3. Figure
4 shows operation at 35V. Output ripple and peak inductor
VSW
20V/DIV
IL
1A/DIV
VOUT
500mV/DIV
(AC COUPLED)
3475 F03
Figure 3. Operation at VIN(MAX) = 20V.
VOUT = 3V and fSW = 1.2MHHz
APPLICATIONS INFORMATION
LT3475/LT3475—1 v w Miummw L7LJCUEN2
LT3475/LT3475-1
10
3475fb
current have signifi cantly increased. Exceeding VIN(MAX)
is safe if the external components have adequate ratings
to handle the peak conditions and if the peak inductor
current does not exceed 3.2A. A saturating inductor may
further reduce performance.
Table 2. Inductors
PART NUMBER
VALUE
(μH)
IRMS
(A)
DCR
()
HEIGHT
(mm)
Sumida
CR43-3R3 3.3 1.44 0.086 3.5
CR43-4R7 4.7 1.15 0.109 3.5
CDRH4D16-3R3 3.3 1.10 0.063 1.8
CDRH4D28-3R3 3.3 1.57 0.049 3.0
CDRH4D28-4R7 4.7 1.32 0.072 3.0
CDRH6D26-5R0 5.0 2.20 0.032 2.8
CDRH6D26-5R6 5.6 2.0 0.036 2.8
CDRH5D28-100 10 1.30 0.048 3.0
CDRH5D28-150 15 1.10 0.076 3.0
CDRH73-100 10 1.68 0.072 3.4
CDRH73-150 15 1.33 0.130 3.4
CDRH104R-150 15 3.1 0.050 4.0
Coilcraft
DO1606T-332 3.3 1.30 0.100 2.0
DO1606T-472 4.7 1.10 0.120 2.0
DO1608C-332 3.3 2.00 0.080 2.9
DO1608C-472 4.7 1.50 0.090 2.9
MOS6020-332 3.3 1.80 0.046 2.0
MOS6020-472 10 1.50 0.050 2.0
DO3316P-103 10 3.9 0.038 5.2
DO3316P-153 15 3.1 0.046 5.2
The optimum inductor for a given application may differ
from the one indicated by this simple design guide. A larger
value inductor provides a higher maximum load current, and
reduces the output voltage ripple. If your load is lower than
the maximum load current, then you can relax the value of the
inductor and operate with higher ripple current. This allows
you to use a physically smaller inductor, or one with a lower
DCR resulting in higher effi ciency. In addition, low inductance
may result in discontinuous mode operation, which further
reduces maximum load current. For details of maximum
output current and discontinuous mode operation, see Linear
Technologys Application Note 44. Finally, for duty cycles
greater than 50% (VOUT/VIN > 0.5), a minimum inductance
is required to avoid sub-harmonic oscillations:
LMIN =(VOUT +VF)800kHz
f
VSW
20V/DIV
IL
1A/DIV
VOUT
500mV/DIV
(AC COUPLED)
3475 F04
Figure 4. Operation above VIN(MAX). Output
Ripple and Peak Inductor Current Increases
Inductor Selection and Maximum Output Current
A good fi rst choice for the inductor value is:
L=(VOUT +VF)1.2MHz
f
where VF is the voltage drop of the catch diode (~0.4V),
f is the switching frequency and L is in μH. With this value
the maximum load current will be above 1.6A at all duty
cycles. The inductors RMS current rating must be greater
than the maximum load current and its saturation current
should be at least 30% higher. For highest effi ciency,
the series resistance (DCR) should be less than 0.15Ω.
Table 2 lists several vendors and types that are suitable.
For robust operation at full load and high input voltages
(VIN > 30V), use an inductor with a saturation current
higher than 3.2A.
APPLICATIONS INFORMATION
LT3475/ LT3475—1 \IVOUTWIN ’Vour) L7 LJUW
LT3475/LT3475-1
11
3475fb
The current in the inductor is a triangle wave with an average
value equal to the load current. The peak switch current
is equal to the output current plus half the peak-to-peak
inductor ripple current. The LT3475 limits its switch cur-
rent in order to protect itself and the system from overload
faults. Therefore, the maximum output current that the
LT3475 will deliver depends on the switch current limit,
the inductor value, and the input and output voltages.
When the switch is off, the potential across the inductor
is the output voltage plus the catch diode drop. This gives
the peak-to-peak ripple current in the inductor
ΔIL=1–DC
()
VOUT +VF
()
L•f
()
where f is the switching frequency of the LT3475 and L
is the value of the inductor. The peak inductor and switch
current is
ISW PK
()
=ILPK
()
=IOUT +ΔIL
2
To maintain output regulation, this peak current must be
less than the LT3475’s switch current limit ILIM. ILIM is at
least 2.3A at low duty cycles and decreases linearly to 1.8A
at DC = 0.9. The maximum output current is a function of
the chosen inductor value:
IOUT MAX
()
= ILIM ΔIL
2
=2.3A• 1–0.25•DC
()
ΔIL
2
Choosing an inductor value so that the ripple current is
small will allow a maximum output current near the switch
current limit.
One approach to choosing the inductor is to start with the
simple rule given above, look at the available inductors,
and choose one to meet cost or space goals. Then use
these equations to check that the LT3475 will be able to
deliver the required output current. Note again that these
equations assume that the inductor current is continu-
ous. Discontinuous operation occurs when IOUT is less
than ΔIL/2.
Input Capacitor Selection
Bypass the input of the LT3475 circuit with a 4.7μF or
higher ceramic capacitor of X7R or X5R type. A lower
value or a less expensive Y5V type will work if there is
additional bypassing provided by bulk electrolytic capaci-
tors or if the input source impedance is low. The following
paragraphs describe the input capacitor considerations in
more detail.
Step-down regulators draw current from the input supply
in pulses with very fast rise and fall times. The input ca-
pacitor is required to reduce the resulting voltage ripple at
the LT3475 input and to force this switching current into a
tight local loop, minimizing EMI. The input capacitor must
have low impedance at the switching frequency to do this
effectively, and it must have an adequate ripple current rat-
ing. With two switchers operating at the same frequency
but with different phases and duty cycles, calculating the
input capacitor RMS current is not simple. However, a
conservative value is the RMS input current for the channel
that is delivering most power (VOUT • IOUT):
CINRMS =IOUT VOUT(V
IN –V
OUT)
VIN
<IOUT
2
and is largest when VIN = 2VOUT (50% duty cycle). As the
second, lower power channel draws input current, the
input capacitors RMS current actually decreases as the
out-of-phase current cancels the current drawn by the
higher power channel. Considering that the maximum
load current from a single channel is ~1.5A, RMS ripple
current will always be less than 0.75A.
The high frequency of the LT3475 reduces the energy
storage requirements of the input capacitor, so that the
capacitance required is less than 10μF. The combination
of small size and low impedance (low equivalent series
resistance or ESR) of ceramic capacitors makes them the
preferred choice. The low ESR results in very low voltage
ripple. Ceramic capacitors can handle larger magnitudes
of ripple current than other capacitor types of the same
value. Use X5R and X7R types.
APPLICATIONS INFORMATION
LT3475/LT3475—1 L7LJCUEN2
LT3475/LT3475-1
12
3475fb
An alternative to a high value ceramic capacitor is a
lower value ceramic along with a larger electrolytic
capacitor. The electrolytic capacitor likely needs to be greater
than 10μF in order to meet the ESR and ripple current
requirements. The input capacitor is likely to see high
surge currents when the input source is applied. Tanta-
lum capacitors can fail due to an over-surge of current.
Only use tantalum capacitors with the appropriate surge
current rating. The manufacturer may also recommend
operation below the rated voltage of the capacitor.
A fi nal caution is in order regarding the use of ceramic
capacitors at the input. A ceramic input capacitor can
combine with stray inductance to form a resonant tank
circuit. If power is applied quickly (for example by plug-
ging the circuit into a live power source) this tank can ring,
doubling the input voltage and damaging the LT3475. The
solution is to either clamp the input voltage or dampen the
tank circuit by adding a lossy capacitor in parallel with the
ceramic capacitor. For details, see Application Note 88.
Output Capacitor Selection
For most LEDs, a 2.2μF, 6.3V ceramic capacitor (X5R or
X7R) at the output results in very low output voltage ripple
and good transient response. Other types and values will
also work. The following discusses tradeoffs in output
ripple and transient performance.
The output capacitor fi lters the inductor current to
generate an output with low voltage ripple. It also stores
energy in order to satisfy transient loads and stabilizes the
LT3475’s control loop. Because the LT3475 operates at a
high frequency, minimal output capacitance is necessary.
In addition, the control loop operates well with or without
the presence of output capacitor series resistance (ESR).
Ceramic capacitors, which achieve very low output ripple
and small circuit size, are therefore an option.
You can estimate output ripple with the following
equation:
V
RIPPLE = ΔIL / (8 • f • COUT) for ceramic capacitors
where ΔIL is the peak-to-peak ripple current in the
inductor. The RMS content of this ripple is very low so the
RMS current rating of the output capacitor is usually not
of concern. It can be estimated with the formula:
IC(RMS) =ΔIL/12
The low ESR and small size of ceramic capacitors make
them the preferred type for LT3475 applications. Not all
ceramic capacitors are the same, however. Many of the
higher value capacitors use poor dielectrics with high
temperature and voltage coeffi cients. In particular Y5V
and Z5U types lose a large fraction of their capacitance
with applied voltage and at temperature extremes.
Because loop stability and transient response depend on
the value of COUT
, this loss may be unacceptable. Use X7R
and X5R types. Table 3 lists several capacitor vendors.
Table 3. Low ESR Surface Mount Capacitors.
VENDOR TYPE SERIES
Taiyo-Yuden Ceramic X5R, X7R
AVX Ceramic X5R, X7R
TDK Ceramic X5R, X7R
Diode Selection
The catch diode (D3 from the Block Diagram) conducts
current only during switch off time. Average forward cur-
rent in normal operation can be calculated from:
I
D(AVG) = IOUT (VIN – VOUT)/VIN
The only reason to consider a diode with a larger current
rating than necessary for nominal operation is for the
worst-case condition of shorted output. The diode cur-
rent will then increase to one half the typical peak switch
current limit.
Peak reverse voltage is equal to the regulator input
voltage. Use a diode with a reverse voltage rating greater
than the input voltage. Table 4 lists several Schottky
diodes and their manufacturers.
Diode reverse leakage can discharge the output capacitor
during LED off times while PWM dimming. If operating at
high ambient temperatures, use a low leakage Schottky
for the widest PWM dimming range.
APPLICATIONS INFORMATION
LT3475/ LT3475—1 L7 LJUW
LT3475/LT3475-1
13
3475fb
Table 4. Schottky Diodes
VR
(V)
IAVE(A)
(A)
VF at 1A
(mV)
VF at 2A
(mV)
On Semiconductor
MBR0540 40 0.5 620
MBRM120E 20 1 530
MBRM140 40 1 550
Diodes Inc
B120 20 1 500
B130 30 1 500
B140HB 40 1 530
DFLS140 40 1.1 510
B240 40 2 500
International Rectifi er
10BQ030 30 1 420
BOOST Pin Considerations
The capacitor and diode tied to the BOOST pin gener-
ate a voltage that is higher than the input voltage. In
most cases, a 0.22μF capacitor and fast switching diode
(such as the CMDSH-3 or MMSD914LT1) will work well.
Figure 5 shows three ways to arrange the boost circuit.
The BOOST pin must be more than 2.5V above the SW
pin for full effi ciency. For outputs of 3.3V and higher, the
standard circuit (Figure 5a) is best. For outputs between
2.8V and 3.3V, use a small Schottky diode (such as the
BAT-54). For lower output voltages, the boost diode can be
tied to the input (Figure 5b). The circuit in Figure 5a is more
effi cient because the BOOST pin current comes from a
lower voltage source. The anode of the boost diode can
be tied to another source that is at least 3V. For example, if
you are generating a 3.3V output, and the 3.3V output is on
whenever the LED is on, the BOOST pin can be
connected to the 3.3V output. For LT3475-1 applications
with higher output voltages, an additional Zener diode
may be necessary (Figure 5d) to maintain pin voltage
below the absolute maximum. In any case, be sure that
the maximum voltage at the BOOST pin is both less than
60V and the voltage difference between the BOOST and
SW pins is less than 30V.
The minimum operating voltage of an LT3475 application
is limited by the undervoltage lockout (~3.7V) and by the
maximum duty cycle. The boost circuit also limits the
minimum input voltage for proper start up. If the input
voltage ramps slowly, or the LT3475 turns on when the
output is already in regulation, the boost capacitor may
not be fully charged. Because the boost capacitor charges
Figure 5. Generating the Boost Voltage
VIN
BOOST
GND
SW
VIN
LT3475
(5a)
D2
VOUT
C3
VBOOST – VSW VOUT
MAX VBOOST VIN + VOUT
VIN
BOOST
GND
SW
VIN
LT3475
(5b)
D2
VOUT
C3
VBOOST – VSW VIN
MAX VBOOST 2VIN
D2
VIN
BOOST
GND
SW
VIN
LT3475
(5c)
3475 F05
VOUT
VBOOST – VSW VIN2
MAX VBOOST VIN2 + VIN
MINIMUM VALUE FOR VIN2 = 3V
VIN2 > 3V
C3
D2
VIN
BOOST
GND
SW
VIN
LT3475
(5d)
3475 F05
VOUT
VBOOST – VSW – VZ
MAX VBOOST VIN + VOUT – VZ
C3
APPLICATIONS INFORMATION
LT3475/LT3475—1 VADJ VADJ j: L7LJCUEN2
LT3475/LT3475-1
14
3475fb
with the energy stored in the inductor, the circuit will rely
on some minimum load current to get the boost circuit
running properly. This minimum load will depend on input
and output voltages, and on the arrangement of the boost
circuit. The minimum load current generally goes to zero
once the circuit has started. The typical performance char-
acteristics section shows a plot of minimum load to start
and to run as a function of input voltage. Even without an
output load current, in many cases the discharged output
capacitor will present a load to the switcher that will allow
it to start. The plots show the worst case, where VIN is
ramping very slowly.
Programming LED Current
The LED current can be set by adjusting the voltage on the
VADJ pin. For a 1.5A LED current, either tie VADJ to REF or
to a 1.25V source. For lower output currents, program the
VADJ using the following formula: ILED = 1.5A • VADJ/1.25V.
Voltages less than 1.25V can be generated with a voltage
divider from the REF pin, as shown in Figure 6. In order
to have accurate LED current, precision resistors are
preferred (1% or better is recommended). Note that the
VADJ pin sources a small amount of bias current, so use
the following formula to choose resistors:
R2 =VADJ
1.25V – VADJ
R1 +50nA
To minimize the error from variations in VADJ pin current,
use resistors with a parallel resistance of less than 4k. Use
resistor strings with a high enough series resistance so as not
to exceed the 500μA current compliance of the REF pin.
Dimming Control
There are several different types of dimming control
circuits. One dimming control circuit (Figure 7) changes
the voltage on the VADJ pin by tying a low on resistance
FET to the resistor divider string. This allows the se-
lection of two different LED currents. For reliable op-
eration program an LED current of no less than 50mA.
The maximum current dimming ratio (IRATIO) can be
calculated from the maximum LED current (IMAX) and the
minimum LED current (IMIN) as follows:
I
MAX/IMIN = IRATIO
Another dimming control circuit (Figure 8) uses the PWM
pin and an external NFET tied to the cathode of the LED.
An external PWM signal is applied to the PWM pin and the
gate of the NFET (For PWM dimming ratios of 20 to 1 or
less, the NFET can be omitted). The average LED current is
proportional to the duty cycle of the PWM signal. When the
PWM signal goes low, the NFET turns off, turning off the
LED and leaving the output capacitor charged. The PWM
pin is pulled low as well, which disconnects the VC pin,
storing the voltage in the capacitor tied there. Use the C-RC
string shown in Figure 8 and Figure 9 tied to the VC pin for
proper operation during startup. When the PWM pin goes
high again, the LED current returns rapidly to its previous
on state since the compensation and output capacitors are
at the correct voltage. This fast settling time allows the
PWM
LED
GND
LT3475
3475 F08
PWM
100Hz TO
10kHz
VC
10k
0.1μF
3.3nF
Figure 8. Dimming Using PWM Signal
REF
VADJ
GND
LT3475
3475 F07
R1
R2
DIM
Figure 7. Dimming with a MOSFET and Resistor Divider
Figure 6. Setting VADJ with a Resistor Divider
REF
VADJ
GND
LT3475
3475 F06
R1
R2
APPLICATIONS INFORMATION
LT3475/ LT3475—1 LTIE®®® ol$>1LM®®® wimmww iTlaeae mefllfllflbflfllfllfl L7 LJUW
LT3475/LT3475-1
15
3475fb
Figure 10. Recommended Component Placement
1
2
3
4
5
6
7
8
9
10
20
19
18
17
16
15
14
13
12
11
3475 F10
VIN
PWM2SHDNPWM1
VIA TO LOCAL GND PLANE
RTVC
GND
LT3475
3475 F09
1M
RT
10k
220pF
0.1μ
F
PWM1
3.3nF
Figure 9. Extending the PWM Dimming Range
LT3475 to maintain diode current regulation with PWM
pulse widths as short as 7.5 switching cycles (12.5μs for
fSW = 600kHz). Maximum PWM period is determined by
the system and is unlikely to be longer than 12ms. Using
PWM periods shorter than 100μs is not recommended.
The maximum PWM dimming ratio (PWMRATIO) can be
calculated from the maximum PWM period (tMAX) and
minimum PWM pulse width (tMIN) as follows:
t
MAX/tMIN = PWMRATIO
Total dimming ratio (DIMRATIO) is the product of the PWM
dimming ratio and the current dimming ratio.
Example:
I
MAX = 1A, IMIN = 0.1A, tMAX = 9.9ms
t
MIN = 3.3μs (fSW = 1.4MHz)
I
RATIO = 1A/0.1A =10:1
PWMRATIO = 9.9ms/3.3μs = 3000:1
DIMRATIO = 10 • 3000 = 30000:1
To achieve the maximum PWM dimming ratio, use the
circuit shown in Figure 9. This allows PWM pulse widths
as short as 4.5 switching cycles (7.5μs for fSW = 600kHz).
Note that if you use the circuit in Figure 9, the rising edge
of the two PWM signals must align within 100ns.
Layout Hints
As with all switching regulators, careful attention must
be paid to the PCB layout and component placement. To
maximize effi ciency, switch rise and fall times are made
as short as possible. To prevent electromagnetic interfer-
ence (EMI) problems, proper layout of the high frequency
switching path is essential. The voltage signal of the SW
and BOOST pins have sharp rise and fall edges. Minimize
the area of all traces connected to the BOOST and SW
pins and always use a ground plane under the switching
regulator to minimize interplane coupling. In addition, the
ground connection for frequency setting resistor RT and
capacitors at VC1, VC2 pins (refer to the Block Diagram)
should be tied directly to the GND pin and not shared
with the power ground path, ensuring a clean, noise-free
connection.
APPLICATIONS INFORMATION
LT3475/LT3475—1 L7LJCUEN2
LT3475/LT3475-1
16
3475fb
BOOST1 BOOST2
SW1
D1
LED 1 LED 2
D2
D3 D4
SW2
OUT1 OUT2
LED1 LED2
VC1 VC2
REF RT
VADJ1 VADJ2
GND
LT3475
VIN SHDN
L2
10μH
L1
10μH
C3
0.22μF
6.3V
C4
0.22μF
6.3V
R2
1k
R3
2k
R1
24.3k
C2
2.2μF
6.3V
C5
2.2μF
6.3V
C6
0.1μF
C7
0.1μF
3475 TA02
C1 TO C5: X5R OR X7R
D1, D2: DFLS140
D3, D4: MBR0540
LED CURRENT = 1A
VIN
5V TO 36V C1
4.7μF
50V
fSW = 600kHz
BOOST1 BOOST2
SW1
LED 11.5A LED
CURRENT
1.5A LED
CURRENT
LED 2
D2
D3 D4
SW2
OUT1 OUT2
LED1 LED2
PWM1 PWM2
VC1 VC2
REF RT
VADJ1 VADJ2
GND
LT3475
VIN SHDN
D1
L2
10μH
L1
10μH
C3
0.22μF
6.3V
C2
0.22μF
6.3V
C4
2.2μF
6.3V
C5
2.2μF
6.3V
C6
3.3nF
C8
0.1μF
C9
0.1μF
C8
220p
C7
3.3nF
3475 TA03
D1, D2: B260
D3, D4: MBR0540
C1 TO C5: X5R OR X7R
M1, M2: Si2302ADS
M3: 2n7002L
VIN
6V TO 36V C1
4.7μF
50V
M1 M2
M3 R1
24.3k
R3
10k
R4
10k
1M
R2
PWM1 PWM2
fSW = 600kHz
Dual Step-Down 1A LED Driver
Dual Step-Down 1.5A LED Driver with 1200 : 1 True Color PWM Dimming
TYPICAL APPLICATIONS
LT3475/ LT3475—1 _I___ .Mr V" 3m LT... LT _I___ Mm H ch L7 LJUW
LT3475/LT3475-1
17
3475fb
BOOST1 BOOST2
SW1
D2
D3 D4
SW2
OUT1 OUT2
LED1
LED2
VC1 VC2
REF RT
VADJ1 VADJ2
GND
LT3475
VIN SHDN
D1
L2
10μH
L1
10μH
C3
0.22μF
6.3V
C2
0.22μF
6.3V
C4
2.2μF
6.3V
C5
2.2μF
6.3V
3475 TA04
D1, D2: B240A
D3, D4: MBR0540
C1 TO C5: X5R OR X7R
VIN
5V TO 36V C1
4.7μF
50V
R1
24.3k
C7
0.1μF
C6
0.1μF
3A LED
CURRENT
LED 1
fSW = 600kHz
BOOST1 BOOST2
SW1
D2
D3 D4
SW2
OUT1
LED1
OUT2
LED2
VC1 VC2
REF RT
VADJ1 VADJ2
GND
LT3475
VIN SHDN
D1
L2
15μH
L1
15μH
C3
0.22μF
10V
C2
0.22μF
10V
C4
2.2μF
10V
C5
2.2μF
10V
3475 TA05
D1, D2: B240A
D3, D4: MMSD4148T1
C1 TO C5: X5R OR X7R
VIN
10V TO 36V C1
4.7μF
50V
R1
24.3k
C7
0.1μF
C6
0.1μF
LED 4LED 3
1.5A LED
CURRENT
LED 2
1.5A LED
CURRENT
LED 1
fSW = 600kHz
Step-Down 3A LED Driver
Dual Step-Down LED Driver with Series Connected LEDs
TYPICAL APPLICATIONS
LT3475/LT3475—1 ul—H— L7LJCUEN2
LT3475/LT3475-1
18
3475fb
BOOST1 BOOST2
SW1
D2
D3
SW2
OUT1
LED1
OUT2
LED2
VC1 VC2
REF RT
VADJ1 VADJ2
GND
LT3475
VIN SHDN
D1
L2
10μH
L1
10μH
C3
0.22μF
35V
C2
0.22μF
35V
C4
2.2μF
6.3V
C5
2.2μF
6.3V
3475 TA06
D1, D2: B240A
D3, D4: MMSD4148T1
C1 TO C5: X5R OR X7R
VIN
5V TO 28V C1
4.7μF
35V
R1
24.3k
C7
0.1μF
C6
0.1μF
LED 2LED 1 1.5A LED
CURRENT
1.5A LED
CURRENT
fSW = 600kHz
D4
Dual Step-Down 1.5A Red LED Driver
TYPICAL APPLICATIONS
LT3475/LT3475—1 iiUJUUUUUUEU i I j flfiflflflflflflffl'j + % {iiifiiifijfj an 7 ‘ + ‘ 14* , 4 ‘ ,,,,, i4 ,7nnnnnnq1qflf @ J J ” “ j HHHHHHHHHH ‘ 4307453‘ ‘ ‘ * ‘ 0099020 4’ E E 01957030 L7 LJUW
LT3475/LT3475-1
19
3475fb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However,
no responsibility is assumed for its use. Linear Technology Corporation makes no representation that
the interconnection of its circuits as described herein will not infringe on existing patent rights.
FE20 (CB) TSSOP 0204
0.09 – 0.20
(.0035 – .0079)
0° 8°
0.25
REF
RECOMMENDED SOLDER PAD LAYOUT
0.50 – 0.75
(.020 – .030)
4.30 – 4.50*
(.169 – .177)
134
5678910
111214 13
6.40 – 6.60*
(.252 – .260)
3.86
(.152)
2.74
(.108)
20 1918 17 16 15
1.20
(.047)
MAX
0.05 – 0.15
(.002 – .006)
0.65
(.0256)
BSC 0.195 – 0.30
(.0077 – .0118)
TYP
2
2.74
(.108)
0.45 ±0.05
0.65 BSC
4.50 ±0.10
6.60 ±0.10
1.05 ±0.10
3.86
(.152)
MILLIMETERS
(INCHES) *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.150mm (.006") PER SIDE
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS
2. DIMENSIONS ARE IN
3. DRAWING NOT TO SCALE
SEE NOTE 4
4. RECOMMENDED MINIMUM PCB METAL SIZE
FOR EXPOSED PAD ATTACHMENT
6.40
(.252)
BSC
FE Package
20-Lead Plastic TSSOP (4.4mm)
(Reference LTC DWG # 05-08-1663)
Exposed Pad Variation CB
PACKAGE DESCRIPTION
LT3475/LT3475—1 L7LJCUEN2
LT3475/LT3475-1
20
3475fb
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 FAX: (408) 434-0507 www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2006
LT 1007 REV B • PRINTED IN USA
RELATED PARTS
PART NUMBER DESCRIPTION COMMENTS
LT1618 Constant-Current, 1.4MHz, 1.5A Boost
Converter
VIN(MIN) = 1.6V, VIN(MAX) = 18V, VOUT(MAX) = 35V, Analog/PWM, ISD < 1μA,
MS10 Package
LT3466 Dual Full Function Step-Up LED Driver Drivers Up to 20 LEDs, VIN: 2.7V to 24V, VOUT(MAX) = 40V, DFN, TSSOP16E Packages
LT3474 36V, 1A (ILED), 2MHz Step-Down
LED Driver
VIN(MIN) = 4V, VIN(MAX) = 36V, 400:1 True Color PWM, ISD < 1μA,
TSSOP16E Package
LT3477 42V, 3A, 3.5MHz Boost, Buck-Boost,
Buck LED Driver
VIN(MIN) = 2.5V, VIN(MAX) = 25V, VOUT(MAX) = 40V, Analog/PWM, ISD < 1μA,
QFN, TSSOP20E Packages
LT3479 3A, Full-Featured DC/DC Converter with
Soft-Start and Inrush Current Protection
VIN(MIN) = 2.5V, VIN(MAX) = 24V, VOUT(MAX) = 40V, Analog/PWM, ISD < 1μA,
DFN, TSSOP Packages
LT3846 Dual 1.3A, 2MHz, LED Driver VIN: 2.5V to 24V, VOUT(MAX) = 36V, 1000:1 True Color PWMTM Dimmin,
DFN, TSSOP16E Packages
BOOST1 BOOST2
SW1 SW2
OUT1 OUT2
LED1
12V TO 18V LED VOLTAGE 12V TO 18V LED VOLTAGE
LED2
VC1 VC2
REF RT
VADJ1 VADJ2
GND
LT3475-1
VIN
VIN
21V TO 36V
SHDN
C2
0.22μF
16V
C1
4.7μF
50V
L1
33μH
R1
1k
R6
100k
R4
10k
R7
100k
R5
10k
L2
33μH
D3
D6
D4
R2
1k
C3
0.22μF
16V
R3
24.3k
C4
2.2μF
25V
C5
2.2μF
25V
D7
Q1
D2D1
D5
C7
0.1μF
C6
0.1μF
3475 TA08
1.5A LED
CURRENT*
1.5A LED
CURRENT*
fSW = 600kHz
D8
Q2
D1, D4: 7.5V ZENER DIODE
D2, D3: MMSD4148
D5, D6: B240A
D7, D8: 22V ZENER DIODE
R1, R2: USE 0.5W RESISTOR OF TWO 2k 0.25W RESISTORS IN PARALLEL
Q1, Q2: MMBT3904
C1 TO C5: X5R or X7R
*DERATE LED CURRENT AT ELEVATED AMBIENT TEMPERATURES TO MAINTAIN LT3475-1 JUNCTION TEMPERATURE BELOW 125 °C.
Dual Step-Down 1.5A LED Driver with Four Series Connected LED Output
TYPICAL APPLICATION

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