MP4460 Datasheet by Monolithic Power Systems Inc.

E9 L1 10w éiz mm 2 2NF v R2 R1 an 2m 124m R3 53.1 m R4 2mm
MP4460
2.5A, 4MHz, 36V
Step-Down Converter
MP4460 Rev. 1.03 www.MonolithicPower.com 1
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The Future of Analog IC Technology
DESCRIPTION
The MP4460 is a high frequency step-down
switching regulator with an integrated internal
high-side high voltage power MOSFET. It
provides 2.5A output with current mode control for
fast loop response and easy compensation.
The wide 3.8V to 36V input range accommodates
a variety of step-down applications, including
those in an automotive input environment. A
120µA operational quiescent current allows use in
battery-powered applications.
High power conversion efficiency over a wide
load range is achieved by scaling down the
switching frequency at light load condition to
reduce the switching and gate driving losses.
The frequency foldback helps prevent inductor
current runaway during startup and thermal
shutdown provides reliable, fault tolerant
operation.
In some applications, such as AM radio and
ADSL applications, in which the device is
sensitive to frequency band, the MP4460 can
avoid the related EMI problem by setting the
frequency at 4MHz.
The MP4460 is available in a small 3mm x 3mm
QFN10 package.
FEATURES
120A Quiescent Current
Wide 3.8V to 36V Operating Input Range
150m Internal Power MOSFET
Up to 4MHz Programmable Switching
Frequency
Ceramic Capacitor Stable
Internal Soft-Start
Internally Set Current Limit without a
Current Sensing Resistor
Up to 95% Efficiency
Output Adjustable from 0.8V to 30V
Available in a 3mm x 3mm QFN10 Package
APPLICATIONS
High Voltage Power Conversion
Automotive Systems
Industrial Power Systems
Distributed Power Systems
Battery Powered Systems
All MPS parts are lead-free and adhere to the RoHS directive. For MPS green
status, please visit MPS website under Products, Quality Assurance page.
“MPS” and “The Future of Analog IC Technology” are registered trademarks of
Monolithic Power Systems, Inc.
TYPICAL APPLICATION
C3
220pF
C4
100nF
D1
VOUT
3.3V
VIN VIN
EN
FREQ
GND
BST SW
FB
COMP
MP4460
EN
6
10
1,2
5
4
8,9
3
7
Efficiency vs
Load Current
100
90
80
70
60
50
40
30
EFFICIENCY (%)
0 0.5 1.0 1.5 2.0 2.5
LOAD CURRENT (A)
V
OUT
=3.3V
V
IN
=12VV
IN
=24V
V
IN
=5V
mp5 MP4460 — 2.5A, 4MHz, 36V STEP-DOWN CONVERTER TJ
MP4460 – 2.5A, 4MHz, 36V STEP-DOWN CONVERTER
MP4460 Rev. 1.03 www.MonolithicPower.com 2
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ORDERING INFORMATION
Part Number* Package Top Marking
MP4460DQ QFN10 (3x3mm) M7
* For Tape & Reel, add suffix –Z (e.g. MP4460DQ–Z);
For RoHS, compliant packaging, add suffix –LF (e.g. MP4460DQ–LF–Z).
PACKAGE REFERENCE
ABSOLUTE MAXIMUM RATINGS (1)
Supply Voltage (VIN).....................–0.3V to +40V
Switch Voltage (VSW)............ –0.3V to VIN + 0.3V
BST to SW .....................................–0.3V to +6V
All Other Pins.................................–0.3V to +6V
Continuous Power Dissipation (TA = +25°C) (2)
QFN10 (3mm x 3mm) ................................ 2.5W
Junction Temperature...............................150C
Lead Temperature ....................................260C
Storage Temperature.............. –65°C to +150C
Recommended Operating Conditions (3)
Supply Voltage VIN ...........................3.8V to 36V
Output Voltage VOUT.........................0.8V to 30V
Operating Temperature........... –40C to +125C
Thermal Resistance (4) θJA θJC
QFN10 (3mm x 3mm) ............. 50 ...... 12... C/W
Notes:
1) Exceeding these ratings may damage the device.
2) The maximum allowable power dissipation is a function of the
maximum junction temperature TJ (MAX), the junction-to-
ambient thermal resistance JA, and the ambient temperature
TA. The maximum allowable continuous power dissipation at
any ambient temperature is calculated by PD (MAX) = (TJ
(MAX)-TA)/JA. Exceeding the maximum allowable powe
r
dissipation will cause excessive die temperature, and the
regulator will go into thermal shutdown. Internal thermal
shutdown circuitry protects the device from permanent
damage.
3) The device is not guaranteed to function outside of its
operating conditions.
4) Measured on JESD51-7, 4-layer PCB
TOP VIEW
SW
SW
EN
COMP
FB
1
2
3
4
5
BST
VIN
VIN
FREQ
GND
10
9
8
7
6
EXPOSED PAD
ON BACKSIDE
mp5 MP4460 — 2.5A, 4MHz, 36V STEP-DOWN CONVERTER VIN UVLO Hysler Soft-Stan Time Mxnimum OffTime Mxnimum On Time
MP4460 – 2.5A, 4MHz, 36V STEP-DOWN CONVERTER
MP4460 Rev. 1.03 www.MonolithicPower.com 3
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ELECTRICAL CHARACTERISTICS
VIN = 12V, VEN = 2.5V, VCOMP = 1.4V, TA= +25C, unless otherwise noted.
Parameter Symbol Condition Min Typ Max Units
Feedback Voltage VFB 4.5V < VIN < 36V 0.776 0.8 0.824 V
Upper Switch On Resistance RDS(ON) VBST – VSW = 5V 150 m
Upper Switch Leakage VEN = 0V, VSW = 0V, VIN = 36V 1 A
Current Limit Duty Cycle = 50% 2.9 3.5 A
COMP to Current Sense
Transconductance GCS 6.3 A/V
Error Amp Voltage Gain (5) 200 V/V
Error Amp Transconductance ICOMP = ±3µA 40 60 80 µA/V
Error Amp Min Source Current VFB = 0.7V 5 µA
Error Amp Min Sink Current VFB = 0.9V –5 µA
VIN UVLO Threshold 2.7 3.0 3.3 V
VIN UVLO Hysteresis 0.35 V
Soft-Start Time (5) 0V < VFB < 0.8V 1.5 ms
RFREQ = 45k 1.6 2 2.4 MHz
Oscillator Frequency RFREQ = 18k 3.2 4 4.8 MHz
Shutdown Supply Current VEN = 0V 12 18 µA
Quiescent Supply Current No load, VFB = 0.9V 120 145 µA
Thermal Shutdown 150 C
Thermal Shutdown Hysteresis 15 C
Minimum Off Time (5) 100 ns
Minimum On Time (5) 100 ns
EN Up Threshold 1.35 1.5 1.65 V
EN Down Threshold 1.15 1.2 1.25 V
Note:
5) Guaranteed by design.
mp5 MP4460 — 2.5A, 4MHz, 36V STEP-DOWN CONVERTER
MP4460 – 2.5A, 4MHz, 36V STEP-DOWN CONVERTER
MP4460 Rev. 1.03 www.MonolithicPower.com 4
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PIN FUNCTIONS
Pin # Name Description
1, 2 SW Switch Node. This is the output from the high-side switch. A low forward drop Schottky diode to
ground is required. The diode must be close to the SW pins to reduce switching spikes.
3 EN
Enable Input. Pulling this pin below the specified threshold shuts the chip down. Pulling it up
above the specified threshold or leaving it floating enables the chip.
4 COMP
Compensation. This node is the output of the error amplifier. Control loop frequency
compensation is applied to this pin.
5 FB
Feedback. This is the input to the error amplifier. The output voltage is set by an resistive
divider connected between the output and GND which scales down VOUT equal to the internal
+0.8V reference.
6 GND
Ground. It should be connected as close as possible to the output capacitor to shorten the high
current switch paths.
7 FREQ
Switching Frequency Program Input. Connect a resistor from this pin to ground to set the
switching frequency.
8, 9 VIN
Input Supply. This supplies power to all the internal control circuitry, both BS regulators and the
high-side switch. A decoupling capacitor to ground must be placed close to this pin to minimize
switching spikes.
10 BST
Bootstrap. This is the positive power supply for the internal floating high-side MOSFET driver.
Connect a bypass capacitor between this pin and SW pin.
l'I'IPj’ Efficiency vs Load Current MP4460 — 2.5A, 4MHz, 36V STEP-DOWN CONVERTER Ef ncy vs Load Current Oscillating Frequency VS- RFREQ mo 100 E 4.0 E 3.5 so an 5 A A no é g E 6 3° 5 3° 3 2.5 E 70 a m E 2.0 E ‘3 w 1 5 u en i so Z w m L? 1.0 j 5° 5° 5 0.5 40 Vour:2 5v VOUT:5V g a a 40 0 05 10 1.5 20 2.5 o 05 1o 15 20 25 10 100 1000 LOAD CURRENT (A) LOAD CURRENT (A) Rpm: (k0) Steady State Steady State Steady State ID“: 01A iQUT:1A low: 2A VDUT VOUT VENT AC Cnupied AC Coupled AC Caupied WW 1UmV/dw zomV/dw . mmvmw -..--__..-_.,~ '1'--'-----~ sz vSW sz . ‘ mV/dw {av/aw L._———..____ 10V/dw a—________ ‘L mew AMM ‘ /\/\/\/\N\/\/\/\/\/ .L W 1A1va 1Ndw 1ms/dw. 2ms/dw fins/div
MP4460 – 2.5A, 4MHz, 36V STEP-DOWN CONVERTER
MP4460 Rev. 1.03 www.MonolithicPower.com 5
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TYPICAL PERFORMANCE CHARACTERISTICS
VIN = 12V, C1 = 10µF, C2 = 22µF, L = 10µH and TA = +25C, unless otherwise noted.
mp5 MP4460 — 2.5A, 4MHz, 36V STEP-DOWN CONVERTER l i 1 W In megs/mu Iouus/dw. {4 WWW/WW W ZOOMS/dw 200us/dw.
MP4460 – 2.5A, 4MHz, 36V STEP-DOWN CONVERTER
MP4460 Rev. 1.03 www.MonolithicPower.com 6
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
VIN = 12V, C1 = 10µF, C2 = 22µF, L = 10µH and TA = +25C, unless otherwise noted.
1ms/div.
Startup
I
OUT
= 0.1A
1ms/div.
Shutdown
I
OUT
= 0.1A
1ms/div.
Startup
I
OUT
= 1A
Shutdown
I
OUT
= 1A
1ms/div.
Startup
I
OUT
= 2A
VOUT
2V/div.
IL
1A/div.
VSW
10V/div.
VEN
5V/div.
VOUT
2V/div.
IL
1A/div.
VSW
10V/div.
VEN
5V/div.
VOUT
2V/div.
IL
1A/div.
VSW
10V/div.
VEN
5V/div.
VOUT
2V/div.
IL
1A/div.
VSW
10V/div.
VEN
5V/div.
VOUT
2V/div.
IL
2A/div.
VSW
10V/div.
VEN
5V/div.
VOUT
2V/div.
IL
1A/div.
Short Circuit Entry
I
OUT
= 0.1A to Short
Short Circuit Recovery
I
OUT
= Short to 0.1A
Shutdown
I
OUT
= 2A
VOUT
2V/div.
IL
1A/div.
VOUT
2V/div.
IL
2A/div.
VSW
10V/div.
VEN
5V/div.
mp5 MP4460 — 2.5A, 4MHz, 36V STEP-DOWN CONVERTER
MP4460 – 2.5A, 4MHz, 36V STEP-DOWN CONVERTER
MP4460 Rev. 1.03 www.MonolithicPower.com 7
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OPERATION
The MP4460 is a variable frequency,
non-synchronous, step-down switching
regulator with an integrated high-side high
voltage power MOSFET. It provides a single
highly efficient solution with current mode
control for fast loop response and easy
compensation. It features a wide input voltage
range, internal soft-start control and precision
current limiting. Its very low operational
quiescent current makes it suitable for battery
powered applications.
PWM Control
At moderate to high output current, the MP4460
operates in a fixed frequency, peak current
control mode to regulate the output voltage. A
PWM cycle is initiated by the internal clock. The
power MOSFET is turned on and remains on
until its current reaches the value set by the
COMP voltage. When the power switch is off, it
remains off for at least 100ns before the next
cycle starts. If, in one PWM period, the current
in the power MOSFET does not reach the
COMP set current value, the power MOSFET
remains on, saving a turn-off operation.
--
+
--
+
1.5ms SS
Gm Error Amp
SS
2.6V
5V
I
SW
COMP
I
SW
VIN
BST
SW
GND FREQ
COMP
FB
EN
SS
0V8
REFERENCE UVLO/
THERMAL
SHUTDOWN
INTERNAL
REGULATORS
OSCILLATOR
--
+
SW
--
+
Level
Shift
CLK
V
OUT
V
IN
V
OUT
Figure 1—Functional Block Diagram
mp5 MP4460 — 2.5A, 4MHz, 36V STEP-DOWN CONVERTER
MP4460 – 2.5A, 4MHz, 36V STEP-DOWN CONVERTER
MP4460 Rev. 1.03 www.MonolithicPower.com 8
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Error Amplifier
The error amplifier compares the FB pin voltage
with the internal reference (REF) and outputs a
current proportional to the difference between
the two. This output current is then used to
charge the external compensation network to
form the COMP voltage, which is used to
control the power MOSFET current.
During operation, the minimum COMP voltage
is clamped to 0.9V and its maximum is clamped
to 2.0V. COMP is internally pulled down to GND
in shutdown mode. COMP should not be pulled
up beyond 2.6V.
Internal Regulator
Most of the internal circuitries are powered from
the 2.6V internal regulator. This regulator takes
the VIN input and operates in the full VIN range.
When VIN is greater than 3.0V, the output of
the regulator is in full regulation. When VIN is
lower than 3.0V, the output decreases.
Enable Control
The MP4460 has a dedicated enable control pin
(EN). With high enough input voltage, the chip
can be enabled and disabled by EN which has
positive logic. Its falling threshold is a precision
1.2V, and its rising threshold is 1.5V (300mV
higher).
When floating, EN is pulled up to about 3.0V by
an internal 1µA current source so it is enabled.
To pull it down, 1µA current capability is needed.
When EN is pulled down below 1.2V, the chip is
put into the lowest shutdown current mode.
When EN is higher than zero but lower than its
rising threshold, the chip is still in shutdown
mode but the shutdown current increases
slightly.
Under-Voltage Lockout (UVLO)
Under-voltage lockout (UVLO) is implemented
to protect the chip from operating at insufficient
supply voltage. The UVLO rising threshold is
about 3.0V while its falling threshold is a
consistent 2.6V.
Internal Soft-Start
The soft-start is implemented to prevent the
converter output voltage from overshooting
during startup. When the chip starts, the
internal circuitry generates a soft-start voltage
(SS) ramping up from 0V to 2.6V. When it is
lower than the internal reference (REF), SS
overrides REF so the error amplifier uses SS as
the reference. When SS is higher than REF,
REF regains control.
Thermal Shutdown
Thermal shutdown is implemented to prevent
the chip from operating at exceedingly high
temperatures. When the silicon die temperature
is higher than its upper threshold, it shuts down
the whole chip. When the temperature is lower
than its lower threshold, the chip is enabled
again.
Floating Driver and Bootstrap Charging
The floating power MOSFET driver is powered
by an external bootstrap capacitor. This floating
driver has its own UVLO protection. This
UVLO’s rising threshold is 2.2V with a threshold
of 150mV.
The bootstrap capacitor is charged and
regulated to about 5V by the dedicated internal
bootstrap regulator. When the voltage between
the BST and SW nodes is lower than its
regulation, a PMOS pass transistor connected
from VIN to BST is turned on. The charging
current path is from VIN, BST and then to SW.
External circuit should provide enough voltage
headroom to facilitate the charging.
As long as VIN is sufficiently higher than SW,
the bootstrap capacitor can be charged. When
the power MOSFET is ON, VIN is about equal
to SW so the bootstrap capacitor cannot be
charged. When the external diode is on, the
difference between VIN and SW is largest, thus
making it the best period to charge. When there
is no current in the inductor, SW equals the
output voltage VOUT so the difference between
VIN and VOUT can be used to charge the
bootstrap capacitor.
mp5 MP4460 — 2.5A, 4MHz, 36V STEP-DOWN CONVERTER
MP4460 – 2.5A, 4MHz, 36V STEP-DOWN CONVERTER
MP4460 Rev. 1.03 www.MonolithicPower.com 9
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At higher duty cycle operation condition, the
time period available to the bootstrap charging
is less so the bootstrap capacitor may not be
sufficiently charged.
In case the internal circuit does not have
sufficient voltage and the bootstrap capacitor is
not charged, extra external circuitry can be
used to ensure the bootstrap voltage is in the
normal operational region. Refer to External
Bootstrap Diode in Application section.
The DC quiescent current of the floating driver
is about 20µA. Make sure the bleeding current
at the SW node is higher than this value, such
that:
A20
)2R1R(
V
IO
O
Current Comparator and Current Limit
The power MOSFET current is accurately
sensed via a current sense MOSFET. It is then
fed to the high speed current comparator for the
current mode control purpose. The current
comparator takes this sensed current as one of
its inputs. When the power MOSFET is turned
on, the comparator is first blanked till the end of
the turn-on transition to avoid noise issues. The
comparator then compares the power switch
current with the COMP voltage. When the
sensed current is higher than the COMP
voltage, the comparator output is low, turning
off the power MOSFET. The cycle-by-cycle
maximum current of the internal power
MOSFET is internally limited.
Startup and Shutdown
If both VIN and EN are higher than their
appropriate thresholds, the chip starts. The
reference block starts first, generating stable
reference voltage and currents, and then the
internal regulator is enabled. The regulator
provides stable supply for the remaining
circuitries.
While the internal supply rail is up, an internal
timer holds the power MOSFET OFF for about
50µs to blank the startup glitches. When the
internal soft-start block is enabled, it first holds
its SS output low to ensure the remaining
circuitries are ready and then slowly ramps up.
Three events can shut down the chip: EN low,
VIN low and thermal shutdown. In the shutdown
procedure, power MOSFET is turned off first to
avoid any fault triggering. The COMP voltage
and the internal supply rail are then pulled down.
Programmable Oscillator
The MP4460 oscillating frequency is set by an
external resistor, RFREQ from the FREQ pin to
ground. The relationship between RFREQ and fS
refer to Table1 in Application section.
mp5 MP4460 — 2.5A, 4MHz, 36V STEP-DOWN CONVERTER fs RFREQ
MP4460 – 2.5A, 4MHz, 36V STEP-DOWN CONVERTER
MP4460 Rev. 1.03 www.MonolithicPower.com 10
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APPLICATION INFORMATION
COMPONENT SELECTION
Setting the Frequency
The MP4460 has an externally adjustable
frequency. The switching frequency (fS) can be
set using a resistor at FREQ pin (RFREQ). The
recommended RFREQ value for various fS, see
Table1.
Table 1—fS vs. RFREQ
RFREQ (k) fS (MHz)
18 4
20 3.8
22.1 3.5
24 3.3
26.7 3
30 2.8
33.2 2.5
39 2.2
45.3 2
51 1.8
57.6 1.6
68 1.4
80.6 1.2
100 1
133 0.8
200 0.5
340 0.3
536 0.2
Setting the Output Voltage
The output voltage is set using a resistive
voltage divider from the output voltage to FB pin.
The voltage divider divides the output voltage
down to the feedback voltage by the
ratio:
2R1R
2R
VV OUTFB
Thus the output voltage is:
2R
)2R1R(
VV FBOUT
About 20µA current from high side BS circuitry
can be seen at the output when the MP4460 is
at no load. In order to absorb this small amount
of current, keep R2 under 40K. A typical
value for R2 can be 40.2k. With this value, R1
can be determined by:
)k)(8.0V(25.501R OUT
For example, for a 3.3V output voltage, R2 is
40.2k, and R1 is 127k.
Inductor
The inductor is required to supply constant
current to the output load while being driven by
the switched input voltage. A larger value
inductor will result in less ripple current that will
result in lower output ripple voltage. However,
the larger value inductor will have a larger
physical size, higher series resistance, and/or
lower saturation current.
A good rule for determining the inductance to
use is to allow the peak-to-peak ripple current in
the inductor to be approximately 30% of the
maximum switch current limit. Also, make sure
that the peak inductor current is below the
maximum switch current limit. The inductance
value can be calculated by:
IN
OUT
LS
OUT
V
V
1
If
V
1L
Where VOUT is the output voltage, VIN is the input
voltage, fS is the switching frequency, and IL is
the peak-to-peak inductor ripple current.
Choose an inductor that will not saturate under
the maximum inductor peak current. The peak
inductor current can be calculated by:
IN
OUT
S
OUT
LOADLP V
V
1
1Lf2
V
II
Where ILOAD is the load current.
Table 2 lists a number of suitable inductors
from various manufacturers. The choice of
which style inductor to use mainly depends on
the price vs. size requirements and any EMI
requirement.
mp5 MP4460 — 2.5A, 4MHz, 36V STEP-DOWN CONVERTER
MP4460 – 2.5A, 4MHz, 36V STEP-DOWN CONVERTER
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Table 2—Inductor Selection Guide
Part Number Inductance (µH) Max DCR () Current Rating (A)
Dimensions
L x W x H (mm3)
Wurth Electronics
7447789002 2.2 0.019 4 7.3x7.3x3.2
7447789003 3.3 0.024 3.42 7.3x7.3x3.2
7447789004 4.7 0.033 2.9 7.3x7.3x3.2
744066100 10 0.035 3.6 10x10x3.8
744771115 15 0.025 3.75 12x12x6
744771122 22 0.031 3.37 12x12x6
TDK
RLF7030T-2R2 2.2 0.012 5.4 7.3x6.8x3.2
RLF7030T-3R3 3.3 0.02 4.1 7.3x6.8x3.2
RLF7030T-4R7 4.7 0.031 3.4 7.3x6.8x3.2
SLF10145T-100 10 0.0364 3 10.1x10.1x4.5
SLF12565T-150M4R2 15 0.0237 4.2 12.5x12.5x6.5
SLF12565T-220M3R5 22 0.0316 3.5 12.5x12.5x6.5
Toko
FDV0630-2R2M 2.2 0.021 5.3 7.7x7x3
FDV0630-3R3M 3.3 0.031 4.3 7.7x7x3
FDV0630-4R7M 4.7 0.049 3.3 7.7x7x3
919AS-100M 10 0.0265 4.3 10.3x10.3x4.5
919AS-160M 16 0.0492 3.3 10.3x10.3x4.5
919AS-220M 22 0.0776 3 10.3x10.3x4.5
Output Rectifier Diode
The output rectifier diode supplies the current to
the inductor when the high-side switch is off. To
reduce losses due to the diode forward voltage
and recovery times, use a Schottky diode.
Choose a diode whose maximum reverse
voltage rating is greater than the maximum
input voltage, and whose current rating is
greater than the maximum load current. Table 3
lists example Schottky diodes and
manufacturers.
Table 3—Diode Selection Guide
Diodes
Voltage/
Current
Rating
Manufacturer
B340A-13-F 40V, 3A Diodes Inc.
CMSH3-40MA 40V, 3A Central Semi
Input Capacitor
The input current to the step-down converter is
discontinuous, therefore a capacitor is required to
supply the AC current to the step-down converter
while maintaining the DC input voltage. Use low
ESR capacitors for the best performance. Ceramic
capacitors are preferred, but tantalum or low-ESR
electrolytic capacitors may also suffice.
For simplification, choose the input capacitor
with RMS current rating greater than half of the
maximum load current.
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MP4460 – 2.5A, 4MHz, 36V STEP-DOWN CONVERTER
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The input capacitor (C1) can be electrolytic,
tantalum or ceramic. When using electrolytic or
tantalum capacitors, a small, high quality
ceramic capacitor, i.e. 0.1F, should be placed
as close to the IC as possible. When using
ceramic capacitors, make sure that they have
enough capacitance to provide sufficient charge
to prevent excessive voltage ripple at input. The
input voltage ripple caused by capacitance can
be estimated by:
IN
OUT
IN
OUT
S
LOAD
IN V
V
1
V
V
1Cf
I
V
Output Capacitor
The output capacitor (C2) is required to
maintain the DC output voltage. Ceramic,
tantalum, or low ESR electrolytic capacitors are
recommended. Low ESR capacitors are
preferred to keep the output voltage ripple low.
The output voltage ripple can be estimated by:
2Cf8
1
R
V
V
1
Lf
V
V
S
ESR
IN
OUT
S
OUT
OUT
Where L is the inductor value and RESR is the
equivalent series resistance (ESR) value of the
output capacitor.
In the case of ceramic capacitors, the
impedance at the switching frequency is
dominated by the capacitance. The output
voltage ripple is mainly caused by the
capacitance. For simplification, the output
voltage ripple can be estimated by:
IN
OUT
2
S
OUT
OUT V
V
1
2CLf8
V
V
In the case of tantalum or electrolytic capacitors,
the ESR dominates the impedance at the
switching frequency. For simplification, the
output ripple can be approximated to:
ESR
IN
OUT
S
OUT
OUT R
V
V
1
Lf
V
V
The characteristics of the output capacitor also
affect the stability of the regulation system. The
MP4460 can be optimized for a wide range of
capacitance and ESR values.
Compensation Components
MP4460 employs current mode control for easy
compensation and fast transient response. The
system stability and transient response are
controlled through the COMP pin. COMP pin is
the output of the internal error amplifier. A
series capacitor-resistor combination sets a
pole-zero combination to control the
characteristics of the control system. The DC
gain of the voltage feedback loop is given by:
OUT
FB
VEACSLOADVDC V
V
AGRA
Where AVEA is the error amplifier voltage gain,
200V/V; GCS is the current sense
transconductance, 3.7A/V; RLOAD is the load
resistor value.
The system has two poles of importance. One
is due to the compensation capacitor (C3), the
output resistor of error amplifier. The other is
due to the output capacitor and the load resistor.
These poles are located at:
VEA
EA
1P A3C2
G
f
LOAD
2P R2C2
1
f
Where, GEA is the error amplifier
transconductance, 60A/V.
The system has one zero of importance, due to
the compensation capacitor (C3) and the
compensation resistor (R3). This zero is located
at:
3R3C2
1
f1Z
The system may have another zero of
importance, if the output capacitor has a large
capacitance and/or a high ESR value. The zero,
due to the ESR and capacitance of the output
capacitor, is located at:
ESR
ESR R2C2
1
f
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In this case (as shown in Figure 2), a third pole
set by the compensation capacitor (C6) and the
compensation resistor (R3) is used to
compensate the effect of the ESR zero on the
loop gain. This pole is located at:
3R6C2
1
f3P
The goal of compensation design is to shape
the converter transfer function to get a desired
loop gain. The system crossover frequency
where the feedback loop has the unity gain is
important. Lower crossover frequencies result
in slower line and load transient responses,
while higher crossover frequencies could cause
system unstable. A good rule of thumb is to set
the crossover frequency to approximately one-
tenth of the switching frequency. The Table 4
lists the typical values of compensation
components for some standard output voltages
with various output capacitors and inductors.
The values of the compensation components
have been optimized for fast transient
responses and good stability at given conditions.
Table 4—Compensation Values for Typical
Output Voltage/Capacitor Combinations
VOUT
(V) L (µH) C2
(µF)
R3
(k)
C3
(pF) C6
1.8 4.7 47 105 100 None
2.5 4.7 - 6.8 22 54.9 220 None
3.3 6.8 -10 22 68.1 220 None
5 15 - 22 22 100 150 None
12 22 - 33 22 147 150 None
To optimize the compensation components for
conditions not listed in Table 3, the following
procedure can be used.
1. Choose the compensation resistor (R3) to set
the desired crossover frequency. Determine the
R3 value by the following equation:
FB
OUT
CSEA
C
V
V
GG
f2C2
3R
Where fC is the desired crossover frequency.
2. Choose the compensation capacitor (C3) to
achieve the desired phase margin. For
applications with typical inductor values, setting
the compensation zero, fZ1, below one forth of
the crossover frequency provides sufficient
phase margin. Determine the C3 value by the
following equation:
C
f3R2
4
3C
3. Determine if the second compensation
capacitor (C6) is required. It is required if the
ESR zero of the output capacitor is located at
less than half of the switching frequency, or the
following relationship is valid:
2
f
R2C2
1S
ESR
If this is the case, then add the second
compensation capacitor (C6) to set the pole fP3
at the location of the ESR zero. Determine the
C6 value by the equation:
3R
R2C
6C ESR
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High Frequency Operation
The switching frequency of MP4460 can be
programmed up to 4MHz by an external resistor.
Please pay attention to the following if the
switching frequency is above 2MHz.
The minimum on time of MP4460 is about 80ns
(typ). Pulse skipping operation can be seen
more easily at higher switching frequency due
to the minimum on time. Recommended
operating voltage is 12V or below, and 24V or
below at 2MHz. Refer to Figure 2 below for
detailed information.
Recommended VIN (max)
vs Switching Frequency
30
25
20
15
10
5
1500 2000 2500 3000 3500 4000
V
OUT
=3.3V
f
s
(KHz)
V
IN (MAX)
(V)
V
OUT
=2.5V
Figure 2—Recommend Max VIN vs. fs
Since the internal bootstrap circuitry has higher
impedance, which may not be adequate to
charge the bootstrap capacitor during each
(1-D)×Ts charging period, an external bootstrap
charging diode is strongly recommended if the
switching frequency is above 2MHz (see
External Bootstrap Diode section for detailed
implementation information).
With higher switching frequencies, the inductive
reactance (XL) of capacitor comes to dominate,
so that the ESL of input/output capacitor
determines the input/output ripple voltage at
higher switching frequency. As a result of that,
high frequency ceramic capacitor is strongly
recommended as input decoupling capacitor
and output filtering capacitor for such high
frequency operation.
Layout becomes more important when the
device switches at higher frequency. It is
essential to place the input decoupling
capacitor, catch diode and the MP4460 (Vin pin,
SW pin and PGND) as close as possible, with
traces that are very short and fairly wide. This
can help to greatly reduce the voltage spike on
SW node, and lower the EMI noise level as well.
Try to run the feedback trace as far from the
inductor and noisy power traces as possible. It
is often a good idea to run the feedback trace
on the side of the PCB opposite of the inductor
with a ground plane separating the two. The
compensation components should be placed
closed to the MP4460. Do not place the
compensation components close to or under
high dv/dt SW node, or inside the high di/dt
power loop. If you have to do so, the proper
ground plane must be in place to isolate those.
Switching loss is expected to be increased at
high switching frequency. To help to improve
the thermal conduction, a grid of thermal vias
can be created right under the exposed pad. It
is recommended that they be small
(15mil barrel diameter) so that the hole is
essentially filled up during the plating process,
thus aiding conduction to the other side. Too
large a hole can cause ‘solder wicking’
problems during the reflow soldering process.
The pitch (distance between the centers) of
several such thermal vias in an area is typically
40mil. Please refer to the layout example on
EV4460 datasheet.
mp5 MP4460 — 2.5A, 4MHz, 36V STEP-DOWN CONVERTER 0.1m: L
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External Bootstrap Diode
It is recommended that an external bootstrap
diode be added when the input voltage is no
greater than 5V or the 5V rail is available in the
system. This helps improve the efficiency of the
regulator. The bootstrap diode can be a low
cost one such as IN4148 or BAT54.
MP4460
SW
BS
5V
Figure 3—External Bootstrap Diode
This diode is also recommended for high duty
cycle operation (when VOUT /VIN >65%) or low
VIN (<5Vin) applications.
At no load or light load, the converter may
operate in pulse skipping mode in order to
maintain the output voltage in regulation. Thus
there is less time to refresh the BS voltage. In
order to have enough gate voltage under such
operating conditions, the difference of VIN –VOUT
should be greater than 3V. For example, if the
VOUT is set to 3.3V, the VIN needs to be higher
than 3.3V+3V=6.3V to maintain enough BS
voltage at no load or light load. To meet this
requirement, EN pin can be used to program
the input UVLO voltage to Vout+3V.
mm MP4460 — 2.5A, 4MHz, 36V STEP-DOWN CONVERTER C4 on H A. 7LHH 3 IV (:2 47 I 63 R1 R2 $40 2m 49 gm R4 __ R3 —— 200m 1 105m V T C4 On I: n O I 02 R5 22% 100m 1 5 3V R4 __ zoom . R3 '
MP4460 – 2.5A, 4MHz, 36V STEP-DOWN CONVERTER
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TYPICAL APPLICATION CIRCUITS
C3
100pF C6
NS
C4
100nF
D1
V
OUT
1.8V
V
IN
6V - 36V
VIN
EN
FREQ
GND
BST SW
FB
COMP
MP4460
EN
6
10
1,2
5
4
8,9
3
7
Figure 4—1.8V Output Typical Application Schematic
C3
150pF C6
NS
C4
100nF
D1
V
OUT
5V
V
IN
10V - 36V
VIN
EN
FREQ
GND
BST SW
FB
COMP
MP4460
EN
6
10
1,2
5
4
8,9
3
7
Figure 5—5V Output Typical Application Schematic
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PCB LAYOUT GUIDE
PCB layout is very important to achieve stable
operation. It is highly recommended to duplicate
EVB layout for optimum performance.
If change is necessary, please follow these
guidelines and take Figure 6 for reference.
1) Keep the path of switching current short
and minimize the loop area formed by Input
cap, high-side MOSFET and external
switching diode.
2) Bypass ceramic capacitors are suggested
to be put close to the VIN Pin.
3) Ensure all feedback connections are short
and direct. Place the feedback resistors
and compensation components as close to
the chip as possible.
4) Route SW away from sensitive analog
areas such as FB.
5) Connect IN, SW, and especially GND
respectively to a large copper area to cool
the chip to improve thermal performance
and long-term reliability.
C1
C3
R2
R1
L1
C4
C2
D1
V
OUT
V
IN
VIN
EN
FREQ
GND
BST SW
FB
COMP
MP4460
EN
R5
R6 R3
R4
MP4460 Typical Application Circuit
TOP Layer Bottom Layer
Figure 6MP4460 Typical Application Circuit and PCB Layout Guide
C4
C1
R2
R3
C3
R1
R5
R4
GNDGND
Vin Vo
SW
L1
C2
SW
EN
FB
D1
SW
COMP
R6
GND
Vin
FREQ
BST
6
7
8
1
2
3
4
5
10
9
Vin
GND
MP4460 — 2.5A, 4MHz, 36V STEP-DOWN CONVERTER l'I'IPj’ Ececc 3.10 Run TYP. Li mwmfim Hmwmm-m
MP4460 – 2.5A, 4MHz, 36V STEP-DOWN CONVERTER
NOTICE: The information in this document is subject to change without notice. Users should warrant and guarantee that third
party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not
assume any legal responsibility for any said applications.
MP4460 Rev. 1.03 www.MonolithicPower.com 18
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PACKAGE INFORMATION
QFN10 (3mm x 3mm)
SIDE VIEW
TOP VIEW
110
65
BOTTOM VIEW
2.90
3.10
1.45
1.75
2.90
3.10
2.25
2.55
0.50
BSC
0.18
0.30
0.80
1.00
0.00
0.05
0.20 REF
PIN 1 ID
MARKING
1.70
0.50
0.25
RECOMMENDED LAND PATTERN
2.90
NOTE:
1) ALL DIMENSIONS ARE IN MILLIMETERS.
2) EXPOSED PADDLE SIZE DOES NOT INCLUDE MOLD FLASH.
3) LEAD COPLANARITY SHALL BE 0.10 MILLIMETER MAX.
4) DRAWING CONFORMS TO JEDEC MO-229, VARIATION VEED-5.
5) DRAWING IS NOT TO SCALE.
PIN 1 ID
SEE DETAIL A
2.50
0.70
PIN 1 ID OPTION B
R0.20 TYP.
PIN 1 ID OPTION A
R0.20 TYP.
DETAIL A
0.30
0.50
PIN 1 ID
INDEX AREA